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MIC2164/-2/-3
Constant Frequency, Synchronous Buck Controllers Featuring Adaptive On-Time Control Hyper Speed ControlTM Family
General Description
The Micrel MIC2164/-2/-3 are constant frequency, synchronous buck controllers featuring adaptive on-time control. The MIC2164/-2/-3 are the first products in the new TM Hyper Speed Control family of buck controllers introduced by Micrel. The MIC2164/-2/-3 controllers operate over an input supply range of 3V to 28V, and are independent of the IC supply voltage. The devices are capable of supplying 25A output current. While the MIC2164 operates at 300kHz, the MIC2164-2 operates at 600kHz, and the MIC2164-3 operates at 1MHz. TM A unique Hyper Speed Control architecture allows for ultrafast transient response while reducing the output capacitance and also makes High VIN/Low VOUT operation possible. The MIC2164/-2/-3 controllers utilizes an architecture which is adaptive Ton ripple controlled. A UVLO feature is provided to ensure proper operation under power-sag conditions to prevent the external power MOSFET from overheating. A soft start feature is provided to reduce the inrush current. Foldback current limit and "hiccup" mode short-circuit protection ensure FET and load protection. The MIC2164/-2/-3 controllers are available in a 10-pin MSOP (MAX1954A-compatible) package with a junction operating range from -40C to +125C. All support documentation can be found on Micrel's web site at: www.micrel.com.
Features
* Hyper Speed Control architecture enables - High delta V operation (VIN=28V and VOUT=0.8V) - Smaller output capacitors than competitors 3V to 28V input voltage Stable with zero-ESR output capacitor 25A output current capability 300kHz/600kHz/1MHz switching frequency Adaptive on-time mode control Output down to 0.8V with 1% FB accuracy Up to 95% efficiency Foldback current limit and "hiccup" mode short-circuit protection 6ms Internal soft start Thermal shutdown Pre-bias output safe -40C to +125C junction temperature range Available in 10-pin MSOP package
TM
* * * * * * * * * * * * *
Applications
* * * * * Set-top box, gateways and routers Printers, scanners, graphic cards and video cards Telecommunication, PCs and servers Microprocessor core supply Low-voltage distributed power
________________________________________________________________________________________________________________________
Typical Application
MIC2164 12V to 3.3V Efficiency
100 95 90
EFFICIENCY (%)
85 80 75 70 65 60 55 50 0 4 8 12 16 20
VIN=5V
MIC2164/-2/-3 Synchronous Controllers Featuring Adaptive On-Time Control
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
OUTPUT CURRENT (A)
Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
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MIC2164/-2/-3
Ordering Information
Part Number MIC2164YMM MIC2164-2YMM MIC2164-3YMM Voltage Adj. Adj. Adj. Switching Frequency 300kHz 600kHz 1MHz Junction Temp. Range -40 to +125C -40 to +125C -40 to +125C Package 10-pin MSOP 10-pin MSOP 10-pin MSOP Lead Finish Pb-Free Pb-Free Pb-Free
Pin Configuration
10-Pin MSOP (MM)
Pin Description
Pin Number 1 Pin Name HSD Pin Function High-Side N-MOSFET Drain Connection (input): Power to the drain of the external high-side N-channel MOSFET. The HSD operating voltage range is from 3V to 28V. Input capacitors between HSD and the power ground (PGND) are required. Enable (input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or floating = enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically 0.8mA). Feedback (input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. Signal ground. GND is the ground path for the device input voltage VIN and the control circuitry. The loop for the signal ground should be separate from the power ground (PGND) loop. Input Voltage (input): Power to the internal reference and control sections of the MIC2164/-2/-3. The IN operating voltage range is from 3V to 5.5V. A 1F and 0.1F ceramic capacitors from IN to GND are recommended for clean operation. Low-Side Drive (output): High-current driver output for external low-side MOSFET. The DL driving voltage swings from ground-to-IN. Power Ground. PGND is the ground path for the MIC2164/-2/-3 buck converter power stage. The PGND pin connects to the sources of low-side N-Channel MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the Signal ground (GND) loop. High-Side Drive (output): High-current driver output for external high-side MOSFET. The DH driving voltage is floating on the switch node voltage (LX). It swings from ground to VIN minus the diode drop. Adding a small resistor between DH pin and the gate of the high-side N-channel MOSFETs can slow down the turn-on and turn-off time of the MOSFETs. Switch Node and Current Sense input: High current output driver return. The LX pin connects directly to the switch node. Due to the high speed switching on this pin, the LX pin should be routed away from sensitive nodes. LX pin also senses the current by monitoring the voltage across the low-side MOSFET during OFF time. In order to sense the current accurately, connect the low-side MOSFET drain to LX using a Kelvin connection. Boost (output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is connected between the IN pin and the BST pin. A boost capacitor of 0.1F is connected between the BST pin and the LX pin. Adding a small resistor in series with the boost capacitor can slow down the turn-on time of high-side N-Channel MOSFETs.
2
EN
3
FB
4 5
GND IN
6 7
DL PGND
8
DH
9
LX
10
BST
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Absolute Maximum Ratings(1)
IN, FB, EN to GND ........................................... -0.3V to +6V BST to LX ......................................................... -0.3V to +6V BST to GND ................................................... -0.3V to +37V DH to LX.............................................-0.3V to (VBST + 0.3V) DL, COMP to GND ...............................-0.3V to (VIN + 0.3V) HSD to GND..................................................... -0.3V to 31V PGND to GND ............................................... -0.3V to +0.3V Junction Temperature .............................................. +150C Storage Temperature (TS)..........................-65C to +150C Lead Temperature (soldering, 10sec) ........................ 260C
Operating Ratings(2)
Input Voltage (VIN) .......................................... 3.0V to 5.5V Supply Voltage (VHSD) ....................................... 3.0V to 28V Operating Temperature Range ..................-40C to +125C Junction Temperature (TJ) .........................-40C to +125C Junction Thermal Resistance MSOP (JA) ............................................130.5C/W Continuous Power Dissipation (TA = 70C)..............421mW (derate 5.6mW/C above 70C)
Electrical Characteristics(4)
VBST - VLX = 5V; TA = 25C, unless noted. Bold values indicate -40C TJ +125C.
Parameter General Operating Input Voltage (VIN) HSD Voltage Range (VHSD) Quiescent Supply Current Standby Supply Current
(6) (5)
Condition
Min
Typ
Max
Units
3.0 3.0 (VFB = 1.5V, output switching but excluding external MOSFET gate current) VIN = VBST = 5.5V, VHSD = 28, LX = unconnected, EN = GND 2.4 1.4 0.8 2.7 50
5.5 28 3.0 2 3
V V mA mA V mV
Under-voltage Lockout Trip Level UVLO Hysteresis DC-DC Controller Output-Voltage Adjust Range (7) (VOUT) Error Amplifier FB Regulation Voltage FB Regulation Voltage FB Input Leakage Current Current-Limit Threshold VFB = 0.8V VFB = 0V Soft-Start Soft-start Period
Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating.
0.8
V
0C TJ 85C -40C TJ 125C
-1 -2 5 103 19 130 48
1 2 500 162 77
% % nA mV mV
6
ms
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 4. Specification for packaged product only. 5. The application is fully functional at low IN (supply of the control section) if the external MOSFETs have enough low voltage VTH. 6. The current will come only from the internal 100k pull-up resistor sitting on the EN Input and tied to IN. 7. The maximum VOUT value is limited by the Fixed TON estimator which obtains VOUT a divided by 6 value (1/6).
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Parameter Oscillator Switching Frequency
(8)
Condition MIC2164 MIC2164-2 MIC2164-3
Min 0.225 0.45 0.75
Typ 0.3 0.6 1 87 74 66 0
Max 0.375 0.75 1.25
Units MHz MHz MHz % % % %
Maximum Duty Cycle
(9)
MIC2164 MIC2164-2 MIC2164-3
Minimum Duty Cycle FET Drives DH, DL Output Low Voltage DH, DL Output High Voltage
Measured at DH
ISINK = 10mA ISOURCE = 10mA VIN-0.1V or VBST-0.1V 2.1 1.8 1.8 1.2 VLX = 28V, VIN = 5.5V,VBST = 33.5V VLX = 28V, VIN = 5.5V,VBST = 33.5V 155 10
0.1
V V
DH On-Resistance, High State DH On-Resistance, Low State DL On-Resistance, High State DL On-Resistance, Low State LX Leakage Current HSD Leakage Current Thermal Protection Over-temperature Shutdown Over-temperature Shutdown Hysteresis Shutdown Control En Logic Level Low En Logic Level High En Pull-up Current
Note: 8. Measured in test mode. 9. Measured at DH. The maximum duty cycle is limited by the fixed mandatory off time Toff of typical 363ns.
3.3 3.3 3.3 2.3 50 20
A A C C
3V < VIN <5.5V 3V < VIN <5.5V
0.4
0.8 0.9 50 1.2
V V A
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Typical Characteristics
100 95 90
MIC2164 12V to 1.5V Efficiency
100 95 90
MIC2164 12V to 3.3V Efficiency
100 95 90
MIC2164-2 12V to 1.5V Efficiency
EFFICIENCY (%)
EFFICIENCY (%)
85 80 75 70 65 60 55 50 0 4 8 12 16 20
85 80 75 70 65 60 55 50 0 4 8 12 16 20
EFFICIENCY (%) VIN=5V
85 80 75 70 65 60 55 50 0 3 6 9 12 15
VIN=5V
VIN=5V
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
100 95 90
MIC2164-2 12V to 3.3V Efficiency
100 95 90
MIC2164-3 12V to 1.5V Efficiency
100 95 90
MIC2164-3 12V to 3.3V Efficiency
EFFICIENCY (%)
EFFICIENCY (%)
85 80 75 70 65 60 55 50 0 3 6 9 12 15
85 80 75 70 65 60 55 50 0 2 4 6 8 10
EFFICIENCY (%)
85 80 75 70 65 60 55 50 0 2 4 6 8 10
VIN=5V
VIN=5V
VIN=5V
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Feedback Voltage vs. Load
0.85 0.84 0.85 0.84
Feedback Voltage vs. Input Voltage
0.85 0.84
Feedback Voltage vs. HSD Voltage
FEEDBACK VOLTAGE (V)
FEEDBACK VOLTAGE (V)
FEEDBACK VOLTAGE (V)
0.83 0.82 0.81 0.80 0.79 0.78 0.77 0.76 0.75 0 4 8 12 16 20
0.83 0.82 0.81 0.80 0.79 0.78 0.77 0.76 0.75 3 3.5 4 4.5 5 5.5
0.83 0.82 0.81 0.80 0.79 0.78 0.77 0.76 0.75 3 8 13 18 23 28
VHSD=12V VIN=5V
VHSD=12V
VIN=5V
OUTPUT CURRENT (A)
INPUT VOLTAGE (V)
HSD VOLTAGE (V)
Feedback Voltage vs. Temperature
0.810
MIC2164 Switching Frequency vs. Load
350 700
MIC2164-2 Switching Frequency vs. Load
SWITCHING FREQUENCY (kHz)
680 660 640 620 600 580 560 540 520 500
0.806 0.804 0.802 0.800 0.798 0.796 0.794 0.792 0.790 -40 -20 0 20 40 60 80 100 120
SWITCHING FREQUENCY (kHz)
0.808
FEEDBACK VOLTAGE (V)
340 330 320 310 300 290 280 270 260 250 0 4 8 12 16 20
VIN=5V
VHSD=12V VIN=5V
VHSD=12V VIN=5V
TEMPERATURE (C)
0
3
6
9
12
15
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
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Typical Characteristics (continued)
MIC2164-3 Switching Frequency vs. Load
1150 350
MIC2164 Switching Frequency vs. VIN
700
MIC2164-2 Switching Frequency vs. VIN
SWITCHING FREQUENCY (kHz)
680 660 640 620 600 580 560 540 520 500 3 3.5 4 4.5 5 5.5
SWITCHING FREQUENCY (kHz)
1090 1060 1030 1000 970 940 910 880 850 0 2 4 6 8 10
SWITCHING FREQUENCY (kHz)
1120
340 330 320 310 300 290 280 270 260 250 3 3.5 4 4.5 5 5.5
VHSD=12V VIN=5V
VHSD=12V
VHSD=12V
OUTPUT CURRENT (A)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
MIC2164-3 Switching Frequency vs. VIN
1150
MIC2164 Switching Frequency vs. VHSD
350 700
MIC2164-2 Switching Frequency vs. VHSD
SWITCHING FREQUENCY (kHz)
680 660 640 620 600 580 560 540 520 500 3 8 13 18 23 28
SWITCHING FREQUENCY (kHz)
1120 1090 1060 1030 1000 970 940 910 880 850 3 3.5 4 4.5 5 5.5
SWITCHING FREQUENCY (kHz)
340 330 320 310 300 290 280 270 260 250 3 8 13 18 23 28
VIN=5V
VIN=5V VOUT=2.5V
VHSD=12V
INPUT VOLTAGE (V)
HSD VOLTAGE (V)
HSD VOLTAGE (V)
MIC2164-3 Switching Frequency vs. VHSD
1150 350
MIC2164 Switching Frequency vs. Temperature
700
MIC2164-2 Switching Frequency vs. Temperature
SWITCHING FREQUENCY (kHz)
680 660 640 620 600 580 560 540 520 500
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
1120 1090 1060 1030 1000 970 940 910 880 850 3 8 13 18 23 28
340 330 320 310 300 290 280 270 260 250 -40 -20 0 20 40 60 80 100 120
VIN=5V
VIN=5V
VIN=5V
-40
-20
0
20
40
60
80
100
120
HSD VOLTAGE (V)
TEMPERATURE (C)
TEMPERATURE (C)
MIC2164-3 Switching Frequency vs. Temperature
1150
150
Current Limit Threshold vs. Feedback Voltage Percentage
150
Current Limit Threshold vs. Temperature
CURRENT LIMIT THRESHOLD (mV)
135 120 105 90 75 60 45 30 15 0 VFB=0.8V VFB=0V
SWITCHING FREQUENCY (kHz)
CURRENT LIMIT THRESHOLD (mV)
1120 1090 1060 1030 1000 970 940 910 880 850 -40 -20 0 20 40 60 80 100 120
135 120 105 90 75 60 45 30 15 0 0 10 20 30 40 50 60 70 80 90 100
VIN=5V
VIN=5V
-40
-20
0
20
40
60
80
100
120
TEMPERATURE (C)
Feedback Voltage Percentage (%)
TEMPERATURE (C)
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Typical Characteristics (continued)
Quiescent Supply Current vs. Input Voltage
2 1.8
QUIESCENT SUPPLY CURRENT (mA)
1.6 1.4 1.2 1 0.8 0.6 0.4 0.2 0 3 3.5 4 4.5 5 5.5
INPUT VOLTAGE (V)
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Functional Characteristics
MIC2164 Switching Waveforms (Light Load)
IL (5A/div) DH (20V/div) LX (10V/div) DL (5V/div)
Vhsd = 12V Vin = 5V Vout = 3.3V L = 1.5H Iout = 1A
Time 2s/div
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Functional Characteristics (continue)
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Functional Characteristics (continue)
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Functional Diagram
Figure 1. MIC2164/-2/-3 Block Diagram
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Functional Description
The MIC2164/-2/-3 is a adaptive on-time synchronous buck controller family built for low cost and high performance. They are designed for wide input voltage range from 3V to 28V and for high output power buck converters. An estimated-ON-time method is applied in MIC2164/-2/-3 to obtain a constant switching frequency and to simplify the control compensation. The overcurrent protection is implemented without the use of an external sense resistor. It includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time. Theory of Operation The MIC2164/-2/-3 is a adaptive on-time buck controller family. Figure 1 illustrates the block diagram for the control loop. The output voltage variation will be sensed by the MIC2164/-2/-3 feedback pin FB via the voltage divider R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low gain transconductance (gm) amplifier, which improves the MIC2164/-2/-3 converter output voltage regulation. If the FB voltage decreases and the output of the gm amplifier is below 0.8V, The error comparator will trigger the control logic and generate an ON-time period, in which DH pin is logic high and DL pin is logic low. The ON-time period length is predetermined by the "FIXED TON ESTIMATION" circuitry:
The estimated-ON-time method results in a constant switching frequency in MIC2164/-2/-3. The actual ON time is varied with the different rising and falling time of the external MOSFETs. Therefore, the type of the external MOSFETs, the output load current, and the control circuitry power supply VIN will modify the actual ON time and the switching frequency. Also, the minimum Ton results in a lower switching frequency in the high VHSD and low VOUT applications, such as 24V to 1.0V MIC2164-3 application. The minimum Ton measured on the MIC2164 evaluation board is about 138ns. During the load transient, the switching frequency is changed due to the varying OFF time. To illustrate the control loop, the steady-state scenario and the load transient scenario are analyzed. For easy analysis, the gain of the gm amplifier is assumed to be 1. With this assumption, the inverting input of the error comparator is the same as the FB voltage. Figure 2 shows the MIC2164/-2/-3 control loop timing during the steady-state. During the steady-state, the gm amplifier senses the FB voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the ON-time period. The ON time is predetermined by the estimation. The ending of OFF time is controlled by the FB voltage. At the valley of the FB voltage ripple, which is below than VREF, OFF period ends and the next ON-time period is triggered through the control logic circuitry.
TON(estimat ed) =
VOUT VHSD fsw
(1)
where VOUT is the output voltage, VHSD is the power stage input voltage, and fSW is the switching frequency (300kHz for MIC2164, 600kHz for MIC2164-2, and 1MHz for MIC2164-3). After ON-time period, the MIC2164/-2/-3 goes into the OFF-time period, in which DH pin is logic low and DL pin is logic high. The OFF-time period length is depending on the FB voltage in most cases. When the FB voltage decreases and the output of the gm amplifier is below 0.8V, the ON-time period is trigger and the OFF-time period ends. If the OFF-time period decided by the FB voltage is less than the minimum OFF time TOFF(min), which is about 363ns typical, the MIC2164/-2/-3 control logic will apply the TOFF(min) instead. TOFF(min) is required by the BST charging. The maximum duty cycle is obtained from the 363ns TOFF(min):
Dmax = TS - TOFF(min) TS = 1- 363ns TS
Figure 2. MIC2164/-2/-3 Control Loop Timing
where Ts = 1/fSW. It is not recommended to use MIC2164/-2/-3 with a OFF time close to TOFF(min) at the steady state.
12
Figure 3 shows the load transient scenario of the MIC2164/-2/-3 converter. The output voltage drops due to the sudden load increasing, which would cause the FB voltage to be less than VREF. This will cause the error comparator to trigger ON-time period. At the end of the ON-time period, a minimum OFF time TOFF(min) is generated to charge BST since the FB voltage is still below the VREF. Then, the next ON-time period is triggered due to the low FB voltage. Therefore, the switching frequency changes during the load transient.
M9999-090409-B
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Micrel, Inc. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in MIC2164/-2/-3 converter.
MIC2164/-2/-3 circuitry is disabled to reduce the current consumption. VIN should be powered up no earlier than VHSD to make the soft-start function behavior correctly.
Current Limit The MIC2164/-2/-3 uses the RDS(ON) of the low-side power MOSFET to sense over-current conditions. The lower-side MOSFET is used because it displays much lower parasitic oscillations during switching then the high-side MOSFET. Using the low-side MOSFET RDS(ON) as a current sense is an excellent method for circuit protection. This method will avoid adding cost, board space and power losses taken by discrete current sense resistors. In each switching cycle of the MIC2164/-2/-3 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. The sensed voltage is compared with a current-limit threshold voltage VCL after a blanking time of 150ns. If the sensed voltage is over VCL, which is 130mV typical at 0.8V feedback voltage, the MIC2164/-2/-3 turns off the high-side MOSFET and a soft-start sequence is trigged. This mode of operation is called the "hiccup mode" and its purpose is to protect the down stream load in case of a hard short. The current limit threshold VCL has a fold back characteristics related to the FB voltage. Please refer to the "Typical Characteristics" for the curve of VCL vs. FB voltage. The circuit in Figure 4 illustrates the MIC2164/-2/-3 current limiting circuit.
Figure 3. MIC2164/-2/-3 Load-Transient Response
Unlike the current-mode control, MIC2164/-2/-3 uses the output voltage ripple, which is proportional to the inductor current ripple if the ESR of the output capacitor is large enough, to trigger an ON-time period. The predetermined ON time makes MIC2164/-2/-3 control loop has the advantage as the adaptive on-time mode control. Therefore, the slope compensation, which is necessary for the current-mode control, is not required in the MIC2164/-2/-3 family. The MIC2164/-2/-3 family has its own stability concern: the FB voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier and the error comparator. The recommended minimum FB voltage ripple is 20mV. If a low ESR output capacitor is selected, the FB voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the FB voltage ripple are not in phase with the inductor current ripple if the ESR of the output capacitor is very low. Therefore, the ripple injection is required for a low ESR output capacitor. Please refer to "Ripple Injection" subsection in "Application Information" for more details about the ripple injection.
Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. MIC2164/-2/-3 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 6ms with a 9.7mV step. Therefore, the output voltage is controlled to increase slowly by a staircase VREF ramp. Once the soft-start ends, the related
Figure 4. MIC2164/-2/-3 Current Limiting Circuit
Using the typical VCL value of 130mV, the current limit value is roughly estimated as:
ICL
130mV R DS(ON)
For designs where the current ripple is significant compared to the load current IOUT, or for low duty cycle operation, calculating the current limit ICL should take into account that one is sensing the peak inductor current and that there is a blanking delay of approximately 150ns.
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ICL
V * TDLY IL(pp) 130mV = + OUT - R DS(ON) L 2 IL(pp) = VOUT (1 - D) f SW L
(2) (3)
where VOUT = The output voltage TDLY = Current limit blanking time, 150ns typical IL(pp) = Inductor current ripple peak-to-peak value D = Duty Cycle fSW = Switching frequency The MOSFET RDS(ON) varies 30% to 40% with temperature; therefore, it is recommended to add a 50% margin to ICL in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect LX pin directly to the drain of the low-side MOSFET to accurately sense the MOSFETs RDS(ON).
MOSFET Gate Drive The MIC2164/-2/-3 high-side drive circuit is designed to switch an N-Channel MOSFET. The Block Diagram of Figure 1 shows a bootstrap circuit, consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the LX pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the LX pin increases to approximately VHSD. Diode D1 is reversed biased and CBST floats high while continuing to keep the highside MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1F to 1F is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. BST = 10mA x 3.33s/0.1F = 333mV for MIC2164. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor RG, which is in series with CBST, can slow down the turn-on time of the high-side N-channel MOSFET. The drive voltage is derived from the supply voltage VIN. The nominal low-side gate drive voltage is VIN and the nominal high-side gate drive voltage is approximately VIN - VDIODE, where VDIODE is the voltage drop across D1. An approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs.
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Application Information
MOSFET Selection The MIC2164/-2/-3 controller works from power stage input voltages of 3V to 28V and has an external 3V to 5.5V VIN to provide power to turn the external N-Channel power MOSFETs for the high- and low-side switches. For applications where VIN < 5V, it is necessary that the power MOSFETs used are sub-logic level and are in full conduction mode for VGS of 2.5V. For applications when VIN > 5V; logic-level MOSFETs, whose operation is specified at VGS = 4.5V must be used. There are different criteria for choosing the high-side and low-side MOSFETs. These differences are more significant at lower duty cycles such as 12V to 1.8V conversion. In such an application, the high-side MOSFET is required to switch as quickly as possible to minimize transition losses, whereas the low-side MOSFET can switch slower, but must handle larger RMS currents. When the duty cycle approaches 50%, the current carrying capability of the high-side MOSFET starts to become critical. It is important to note that the on-resistance of a MOSFET increases with increasing temperature. A 75C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current limit. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2164/-2/-3 gate-drive circuit. At 300kHz switching frequency and above, the gate charge can be a significant source of power dissipation in the MIC2164/2/-3. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is:
For the low-side MOSFET: IG[low - side] (avg) = C ISS x VGS x f SW (5) Since the current from the gate drive comes from the VIN, the power dissipated in the MIC2164/-2/-3 due to gate drive is: PGATEDRIVE = VIN .(IG[high- side] (avg) + IG[low -side] (avg)) (6) A convenient figure of merit for switching MOSFETs is the on resistance times the total gate charge RDS(ON) x QG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2164/-2/-3. Also, the RDS(ON) of the lowside MOSFET will determine the current limit value. Please refer to "Current Limit" subsection is "Functional Description" for more details. Parameters that are important to MOSFET switch selection are: * * * Voltage rating On-resistance
Total gate charge The voltage ratings for the high-side and low-side MOSFETs are essentially equal to the power stage input voltage VHSD. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitic elements. The power dissipated in the MOSFETs is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses during the period of time when the MOSFETs turn on and off (PAC). PSW = PCONDUCTION + PAC
PCONDUCTION = I SW(RMS) * R DS(ON) PAC = PAC(off ) + PAC(on)
2
(7)
(8) (9)
IG[high-side] (avg) = Q G x f SW
(4)
where: IG[high-side](avg) = Average high-side MOSFET gate current QG = Total gate charge for the high-side MOSFET taken from the manufacturer's data sheet for VGS = VIN. fSW = Switching Frequency The low-side MOSFET is turned on and off at VDS = 0 because an internal body diode or external freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge.
where: RDS(ON) = on-resistance of the MOSFET switch D = Duty Cycle = VOUT / VHSD Making the assumption that the turn-on and turn-off transition times are equal; the transition times can be approximated by: tT = C ISS x VIN + C OSS x VHSD IG (10)
where: CISS and COSS are measured at VDS = 0 IG = gate-drive current The total high-side MOSFET switching loss is: PAC = (VHSD + VD ) x IPK x t T x f SW (11)
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Micrel, Inc. where: tT = Switching transition time VD = Body diode drop (0.5v) fSW = Switching Frequency The high-side MOSFET switching losses increase with the switching frequency and the input voltage VHSD. The low-side MOSFET switching losses are negligible and can be ignored for these calculations.
Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below.
MIC2164/-2/-3
Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below: 2 PINDUCTORCu=IL(RMS) x RWINDING (16) The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. RWINDING = RWINDING(20c) x (1 + 0.0042 x (TH - T20C)) (17) where: TH = temperature of wire under full load T20C = ambient temperature RWINDING(20C) = room temperature winding resistance (usually specified by the manufacturer)
Output Capacitor Selection The type of the output capacitor is usually determined by its ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitors are tantalum, low-ESR aluminum electrolytic, OS-CON and POSCAPS. The output capacitor's ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR is calculated:
L=
VOUT x VHSD(max) - VOUT
(
)
VHSD(max) x f SW x 20% x IOUT(max)
(12)
where: fSW = switching frequency 20% = ratio of AC ripple current to DC output current VHSD(max) = maximum power stage input voltage The peak-to-peak inductor current ripple is: IL(PP ) = VOUT x ( VHSD(max) - VOUT ) VHSD(max) x f SW x L (13)
ESR COUT
VOUT(pp) IL(PP)
(18)
The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(PK) = IOUT(max) + 0.5 x IL(PP) (14)
2
The RMS inductor current is used to calculate the I R losses in the inductor.
where: VOUT(pp) = peak-to-peak output voltage ripple IL(PP) = peak-to-peak inductor current ripple The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated below:
IL(PP) 2 + IL(PP) ESR C VOUT(pp) = OUT C f SW 8 OUT (19)
2
IL(RMS) = IOUT(max)2 +
IL(PP) 12
2
(15)
(
)
Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2164/-2/-3 requires the use of ferrite materials for all but the most cost sensitive applications.
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Micrel, Inc. Where: D = duty cycle COUT = output capacitance value fSW = switching frequency As described in the "Theory of Operation" subsection in "Functional Description", MIC2164/-2/-3 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator to behavior properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitor COUT should be much smaller than the ripple caused by the output capacitor ESR. If low ESR capacitors are selected as the output capacitors, such as ceramic capacitors, a ripple injection method is applied to provide the enough FB voltage ripples. Please refer to the "Ripple Injection" subsection for more details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated below: ICOUT (RMS) = IL(PP) 12
2
MIC2164/-2/-3
output voltage, as shown in Figure 5.
Figure 5. Voltage-Divider Configuration
The output voltage is determined by the equation: R1 ) (25) R2 where VREF = 0.8V. A typical value of R1 can be between 3k and 10k. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: VOUT = VREF (1 + R2 = VREF R1 VOUT - VREF (26)
(20)
The power dissipated in the output capacitor is:
PDISS(COUT ) = I COUT (RMS) ESR COUT
(21)
Input Capacitor Selection The input capacitor for the power stage input VHSD should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor's voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. The input voltage ripple will primarily depend upon the input capacitor's ESR. The peak input current is equal to the peak inductor current, so:
External Schottky Diode (Optional) An external freewheeling diode, which is not necessary, can be used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead time prevents current from flowing unimpeded through both MOSFETs and is typically 30ns. The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current.
ID(avg) = I OUT 2 30ns f SW The reverse voltage requirement of the diode is: VDIODE(rrm) = VHSD The power dissipated by the Schottky diode is: PDIODE = ID(avg) x VF
(27)
VIN = IL(PK ) x ESR CIN
(22)
(28)
The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: ICIN(RMS) IOUT(max) x D x (1 - D ) The power dissipated in the input capacitor is: 2 PDISS(CIN) = ICIN(RMS) xESRCIN (23) (24)
Voltage Setting Components The MIC2164/-2/-3 requires two resistors to set the
where, VF = forward voltage at the peak diode current. The external Schottky diode is not necessary for the circuit operation since the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease the high frequency noise. If the MOSFET body diode is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode
17
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Micrel, Inc. becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the circuit inductance will cause ringing during the high-side MOSFET turn-on. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending upon the circuit components and operating conditions, an external Schottky diode will give a 1/2% to 1% improvement in efficiency.
Ripple Injection The minimum FB voltage ripple requested by the MIC2164/-2/-3 gm amplifier and error comparator is 20mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as 1V output, the output voltage ripple is only 10mV to 20mV, and the FB voltage ripple is less than 20mV. If the FB voltage ripple is so small that the gm amplifier and error comparator could not sense it, the MIC2164/-2/-3 will lose control and the output voltage is not regulated. In order to have some amount of FB voltage ripple, the ripple injection method is applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the FB voltage ripple: 1) Enough ripple at the FB voltage due to the large ESR of the output capacitors. As shown in Figure 6a, the converter is stable without any adding in this situation. The FB voltage ripple is:
MIC2164/-2/-3
Figure 6a. Enough Ripple at FB
Figure 6b. Inadequate Ripple at FB
Figure 6c. Invisible Ripple at FB
VFB(pp) =
where IL(pp) current ripple. 2) Inadequate ripple at the FB voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feedforward capacitor Cff in this situation, as shown in Figure 6b. The typical Cff value is between 1nF to 100nF. With the feedforward capacitor, the FB voltage ripple is very close to the output voltage ripple: VFB(pp) ESR IL (pp) (30) 3) Invisible ripple at the FB voltage due to the very low ESR of the output capacitors.
R2 ESR COUT IL (pp) (29) R1 + R2 is the peak-to-peak value of the inductor
In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node LX via a resistor Rinj and a capacitor Cinj, as shown in Figure 6c. The injected ripple is: 1 VFB(pp) = VHSD x K div x D x (1- D) x (31) f SW x
K div = R1//R2 Rinj + R1//R2 (32)
where VHSD = Power stage input voltage at HSD pin D = Duty Cycle fSW = switching frequency
= (R1// R2 // Rinj) Cff
In the formula (31) and (32), it is assumed that the time constant associated with Cff must be much greater than the switching period:
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Step 2. Select Rinj according to the expected feedback voltage ripple. According to the equation (32),
1 T = << 1 fsw x
If the voltage divider resistors R1 and R2 are in the k range, a Cff of 1nF to 100nF can easily satisfy the large time constant consumption. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in k range.
K div =
VFB(pp )
VHSD
f SW D (1 - D) 1 K div
(33)
Then the value of Rinj is obtained as: R inj = (R1 // R2) (
- 1)
(34)
Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies.
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MIC2164/-2/-3 capacitor is very critical. Connections must be made with wide trace.
Inductor
PCB Layout Guideline
Warning!!! To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC2164/-2/-3 converter. IC
* * * *
Keep the inductor connection to the switch node (LX) short. Do not route any digital lines underneath or close to the inductor. Keep the switch node (LX) away from the feedback (FB) pin. The LX pin should be connected directly to the drain of the low-side MOSFET to accurate sense the voltage across the low-side MOSFET. To minimize noise, place a ground plane underneath the inductor. Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation. Place the Schottky diode on the same side of the board as the MOSFETs and HSD input capacitor. The connection from the Schottky diode's Anode to the input capacitors ground terminal must be as short as possible. The diode's Cathode connection to the switch node (LX) must be keep as short as possible. Place the RC snubber on the same side of the board and as close to the MOSFETs as possible.
* * *
Place the IC and MOSFETs close to the point of load (POL). Use fat traces to route the input and output power lines. Signal and power grounds should be kept separate and connected at only one location. Place the HSD input capacitor next. Place the HSD input capacitors on the same side of the board and as close to the MOSFETs as possible. Keep both the HSD and PGND connections short. Place several vias to the ground plane close to the HSD input capacitor ground terminal. Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. In "Hot-Plug" applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. An additional Tantalum or Electrolytic bypass input capacitor of 22uF or higher is required at the input power connection. The 1F and 0.1F capacitors, which connect to the VIN terminal, must be located right at the IC. The VIN terminal is very noise sensitive and placement of the
*
Output Capacitor
*
Input Capacitor
* *
*
* * * *
*
Schottky Diode (Optional)
* *
*
*
*
RC Snubber
*
*
*
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Evaluation Board Schematics
Figure 7. Schematic of MIC2164 20A Evaluation Board
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MIC2164/-2/-3
Bill of Materials
Item C1, C8, C17, C19 C2 C3 C4, C5, C6 C9 C10 C11 C12 C15 D1 L1 Q1 Q2, Q3 Q5 R1 R2, R7 R3 R5 R6, R9 R12 R13 R14 R15 U1 U2 Part Number 06035C104KAT 0805ZD225MAT GRM216R61A225ME24D C2012X5R1A225K/0.85 222215095001 1210YD226MAT GRM32ER61C226ME20L 0805ZD105KAT GRM219R61A105MA01D 06035C223KAT GRM188R71H223MA01D 16ME1000WGL 12106D107MAT GRM32ER60J107ME20L C3225X5R0J107M 06035C102KAT SD103BWS CDEP147NP-1R5M FDS6699S FDS7766S CMPDM7002A 2N7002E-T1-E3 CRCW06032R21FKEY3 CRCW06030000FKEY3 CRCW06034992FKEY3 CRCW06031R21FKEY3 CRCW06031002FKEY3 CRCW06034751FKEY3 CRCW06038061FKEY3 CRCW06034022FKEY3 CRCW06033241FKEY3 MIC2164YMM MIC5233-5.0YM5 Manufacturer AVX
(1)
Description 0.1F Ceramic Capacitor, X7R, Size 0603, 50V 2.2F Ceramic Capacitor, X5R, Size 0805, 10V 220F Aluminum Capacitor, SMD, 35V 22F Ceramic Capacitor, X5R, Size 1210, 16V 1F Ceramic Capacitor, X5R, Size 0805, 10V 22nF Ceramic Capacitor, X7R, Size 0603, 50V 22nF Ceramic Capacitor, X7R, 0603, 50V 1000F Aluminum Capacitor, 16V 100F Ceramic Capacitor, X5R, Size 1210, 6.3V 1nF Ceramic Capacitor, X7R, 0603, 50V Small Signal Schottky Diode 1.5H Inductor, 27.2A Saturation Current 30V N-Channel MOSFET 4.5m Rds(on) @ 4.5V 30V N-Channel MOSFET 6.5m Rds(on) @ 4.5V
(8)
Qty 4 1 1 3 1 1 1 1 1 1 1 1 2 1 1 2 1 1 2 1 1 1 1 1 1
AVX MuRata TDK Vishay AVX MuRata AVX MuRata AVX MuRata Sanyo AVX muRata TDK AVX Vishay Sumida
(6) (7) (5) (2) (3) (4)
Fairchild
Fairchild Central Semiconductor Vishay Vishay-Dale
(4)
Signal MOSFET, 60V 2.21 Resistor, Size 0603, 1% 0 Resistor, Size 0603, 1% 49.9k Resistor, Size 0603, 1% 1.21 Resistor, Size 0603, 1% 10k Resistor, Size 0603, 1% 4.75k Resistor, Size 0603, 1% 8.06k Resistor, Size 0603, 1% 40.2k Resistor, Size 0603, 1% 3.24k Resistor, Size 0603, 1% 300kHz Buck Controller LDO
Vishay-Dale Vishay-Dale Vishay-Dale Vishay-Dale Vishay-Dale Vishay-Dale Vishay-Dale Vishay-Dale Micrel. Inc.
(9)
Micrel. Inc.
September 2009
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Notes: 1. 2. 3. 4. 5. 6. 7. 8. 9. AVX: www.avx.com MuRata: www.murata.com TDK: www.tdk.com Vishay: www.vishay.com Sanyo: www.sanyo.com Sumida: www.sumida.com Fairchild: www.fairchildsemi.com Central Semiconductor: www.centralsemi.com Micrel, Inc: www.micrel.com
MIC2164/-2/-3
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MIC2164/-2/-3
PCB Layout
Figure 8. MIC2164 20A Evaluation Board Top Layer
Figure 9. MIC2164 20A Evaluation Board Bottom Layer
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Figure 10. MIC2164 20A Evaluation Board Mid-Layer 1
Figure 11. MIC2164 20A Evaluation Board Mid-Layer 2
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Application Schematics and Bill of Materials
Figure 12. MIC2164 12V to 3.3V @ 20A Buck Converter
Bill of Materials (MIC2164 12V to 3.3V @ 20A)
Item C1, C8, C17, C19 C2 C3 C4, C5, C6 C9 C10 C11 C12 C15 D1 L1 Q1 Part Number 06035C104KAT 0805ZD225MAT 222215095001 1210YD226MAT 0805ZD105KAT 06035C223KAT 16ME1000WGL 12106D107MAT 06035C102KAT SD103BWS CDEP147NP-1R5M FDS6699S Manufacturer AVX
(1)
Description 0.1F Ceramic Capacitor, X7R, Size 0603, 50V 2.2F Ceramic Capacitor, X5R, Size 0805, 10V 220F Aluminum Capacitor, SMD, 35V 22F Ceramic Capacitor, X5R, Size 1210, 16V 1F Ceramic Capacitor, X5R, Size 0805, 10V 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1000F Aluminum Capacitor, 16V 100F Ceramic Capacitor, X5R, Size 1210, 6.3V 1nF Ceramic Capacitor, X7R, Size 0603, 50V Small Signal Schottky Diode 1.5H Inductor, 27.2A Saturation Current 30V N-Channel MOSFET 4.5m Rds(on) @ 4.5V
Qty 4 1 1 3 1 1 1 1 1 1 1 1
AVX Vishay AVX AVX AVX Sanyo
(3) (2)
AVX AVX Vishay Sumida
(4) (5)
Fairchild
September 2009
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MIC2164/-2/-3
Bill of Materials (MIC2164 12V to 3.3V @ 20A)
Item Q2, Q3 R1 R5 R6, R9 R15 U1 U2
Notes: 1. 2. 3. 4. 5. 6. AVX: www.avx.com Vishay: www.vishay.com Sanyo: www.sanyo.com Sumida: www.sumida.com Fairchild: www.fairchildsemi.com Micrel, Inc: www.micrel.com.
Part Number FDS7766S CRCW06032R21FKEY3 CRCW06031R21FKEY3 CRCW06031002FKEY3 CRCW06033241FKEY3 MIC2164YMM MIC5233-5.0YM5
Manufacturer Fairchild Vishay Dale Vishay Dale Vishay Dale Vishay Dale Micrel. Inc.
(6)
Description 30V N-Channel MOSFET 6.5m Rds(on) @ 4.5V 2.21 Resistor, Size 0603, 1% 1.21 Resistor, Size 0603, 1% 10k Resistor, Size 0603, 1% 3.24k Resistor, Size 0603 1% 300kHz Buck Controller LDO
Qty 2 1 1 2 1 1 1
Micrel. Inc.
September 2009
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MIC2164/-2/-3
Figure 13. MIC2164 12V to 1.8V @ 10A Buck Converter
Bill of Materials (MIC2164 12V to 1.8V @ 10A)
Item C1, C8, C17, C19 C2 C3 C4, C5 C9 C10 C11 C12 C15 D1 L1 Q1, Q2 Part Number 06035C104KAT 0805ZD225MAT 222215095001 1210YD106MAT 0805ZD105KAT 06035C223KAT 6SEPC560MX 12106D107MAT 06035C102KAT SD103BWS CDEP105-2R0MC-32 FDS7764A Manufacturer AVX
(1)
Description 0.1F Ceramic Capacitor, X7R, Size 0603, 50V 2.2F Ceramic Capacitor, X5R, Size 0805, 10V 220F Aluminum Capacitor, SMD, 35V 10F Ceramic Capacitor, X5R, Size 1210, 16V 1F Ceramic Capacitor, X5R, Size 0805, 10V 22nF Ceramic Capacitor, X7R, Size 0603, 50V 560F OSCON Capacitor, 6.3V 100F Ceramic Capacitor, X5R, Size 1210, 6.3V 1nF Ceramic Capacitor, X7R, Size 0603, 50V Small Signal Schottky Diode 2.0H Inductor, 15.8A Saturation Current 30V N-Channel MOSFET 7.5m Rds(on) @ 4.5V
Qty 4 1 1 2 1 1 1 1 1 1 1 2
AVX Vishay AVX AVX AVX Sanyo
(3) (2)
AVX AVX Vishay Sumida
(4) (5)
Fairchild
September 2009
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MIC2164/-2/-3
Bill of Materials (MIC2164 12V to 1.8V @ 10A)
Item R1 R5 R6, R9 R15 U1 U2
Notes: 1. 2. 3. 4. AVX: www.avx.com Vishay: www.vishay.com Sanyo: www.sanyo.com Sumida: www.sumida.com Fairchild: www.fairchildsemi.com Micrel, Inc: www.micrel.com.
Part Number CRCW06032R21FKEY3 CRCW06031R21FKEY3 CRCW06031002FKEY3 CRCW06038061FKEY3 MIC2164YMM MIC5233-5.0YM5
Manufacturer Vishay Dale Vishay Dale Vishay Dale Vishay Dale Micrel. Inc.
(6)
Description 2.21 Resistor, Size 0603, 1% 1.21 Resistor, Size 0603, 1% 10k Resistor, Size 0603, 1% 8.06k Resistor, Size 0603, 1% 300kHz Buck Controller LDO
Qty 1 1 2 1 1 1
Micrel. Inc.
5. 6.
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Figure 14. MIC2164 12V to 1.0V @ 5A Buck Converter
Bill of Materials (MIC2164 12V to 1.0V @ 5A)
Item C1, C8, C17, C19 C2 C3 C4 C9 C10 C11, C12, C13 C15 D1 L1 Q1 R1 R5 Part Number 06035C104KAT 0805ZD225MAT 222215095001 1210YD106MAT 0805ZD105KAT 06035C223KAT 12106D107MAT 06035C102KAT SD103BWS CDRH104RNP-3R8 FDS6910 CRCW06032R21FKEY3 CRCW06031R21FKEY3 Manufacturer AVX
(1)
Description 0.1F Ceramic Capacitor, X7R, Size 0603, 50V 2.2F Ceramic Capacitor, X5R, Size 0805, 10V 220F Aluminum Capacitor, SMD, 35V 10F Ceramic Capacitor, X5R, Size 1210, 16V 1F Ceramic Capacitor, X5R, Size 0805, 10V 22nF Ceramic Capacitor, X7R, Size 0603, 50V 100F Ceramic Capacitor, X5R, Size 1210, 6.3V 1nF Ceramic Capacitor, X7R, Size 0603, 50V Small Signal Schottky Diode 3.8H Inductor, 6A Saturation Current Dual 30V N-Channel MOSFET 17m Rds(on) @ 4.5V 2.21 Resistor, Size 0603, 1% 1.21 Resistor, Size 0603, 1%
Qty 4 1 1 1 1 1 3 1 1 1 1 1 1
AVX Vishay AVX AVX AVX AVX AVX Vishay Sumida
(4) (5) (2) (2)
Fairchild
Vishay Dale
Vishay Dale
September 2009
30
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164 12V to 1.0V @ 5A)
Item R6, R9 R15 U1 U2
Notes: 1. 2. 3. 4. 5. AVX: www.avx.com Vishay: www.vishay.com Sanyo: www.sanyo.com Sumida: www.sumida.com Fairchild: www.fairchildsemi.com Micrel, Inc: www.micrel.com
Part Number CRCW06031002FKEY3 CRCW06034022FKEY3 MIC2164YMM MIC5233-5.0YM5
Manufacturer Vishay Dale Vishay Dale Micrel. Inc.
(6)
Description 10k Resistor, Size 0603, 1% 40.2k Resistor, Size 0603, 1% 300kHz Buck Controller LDO
Qty 2 1 1 1
Micrel. Inc.
6.
September 2009
31
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Figure 15. MIC2164-2 12V to 3.3V @ 15A Buck Converter
Bill of Materials (MIC2164-2 12V to 3.3V @ 15A)
Item C1, C8, C17, C19 C2 C3 C4, C5 C9 C10 C11 C12 C15 D1 L1 Q1 Part Number 06035C104KAT 0805ZD225MAT 222215095001 1210YD226MAT 0805ZD105KAT 06035C472KAT 16ME1000WGL 12106D107MAT 06035C102KAT SD103BWS HCP1305-1R0 FDS8672S Manufacturer AVX
(1)
Description 0.1F Ceramic Capacitor, X7R, Size 0603, 50V 2.2F Ceramic Capacitor, X5R, Size 0805, 10V 220F Aluminum Capacitor, SMD, 35V 22F Ceramic Capacitor, X5R, Size 1210, 16V 1F Ceramic Capacitor, X5R, Size 0805, 10V 4.7nF Ceramic Capacitor, X7R, Size 0603, 50V 1000F Aluminum Capacitor, 16V 100F Ceramic Capacitor, X5R, Size 1210, 6.3V 1nF Ceramic Capacitor, X7R, Size 0603, 50V Small Signal Schottky Diode 1.0H Inductor, 29A DC Current 30V N-Channel MOSFET 7.0m Rds(on) @ 4.5V
Qty 4 1 1 2 1 1 1 1 1 1 1 1
AVX Vishay AVX AVX AVX Sanyo
(3) (2)
AVX AVX Vishay Cooper (4) Bussmann Fairchild
(5)
September 2009
32
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164-2 12V to 3.3V @ 15A)
Item Q2, Q3 R1 R5 R6 R6 R15 U1 U2
Notes: 1. 2. 3. 4. 5. 6. AVX: www.avx.com Vishay: www.vishay.com Sanyo: www.sanyo.com Cooper Bussmann: www.cooperbussmann.com Fairchild: www.fairchildsemi.com Micrel, Inc: www.micrel.com.
Part Number FDS8874 CRCW06032R21FKEY3 CRCW06031R21FKEY3 CRCW06031002FKEY3 CRCW06034021FKEY3 CRCW06033241FKEY3 MIC2164-2YMM MIC5233-5.0YM5
Manufacturer Fairchild Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Micrel. Inc.
(6)
Description 30V N-Channel MOSFET 7.0m Rds(on) @ 4.5V 2.21 Resistor, Size 0603, 1% 1.21 Resistor, Size 0603, 1% 10k Resistor, Size 0603, 1% 4.02k Resistor, Size 0603, 1% 3.24k Resistor, Size 0603 1% 600kHz Buck Controller LDO
Qty 2 1 1 1 1 1 1 1
Micrel. Inc.
September 2009
33
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Figure 16. MIC2164-3 12V to 1.8V @ 10A Buck Converter
Bill of Materials (MIC2164-3 12V to 1.8V @ 10A)
Item C1, C8, C17, C19 C2 C3 C4 C9 C10 C11 C12 C15 D1 L1 Q1, Q2 R1 Part Number 06035C104KAT 0805ZD225MAT 222215095001 1210YD106MAT 0805ZD105KAT 06035C222KAT 6SEPC560MX 12106D107MAT 06035C102KAT SD103BWS HCF1305-1R0 FDS8672S CRCW06032R21FKEY3 Manufacturer AVX
(1)
Description 0.1F Ceramic Capacitor, X7R, Size 0603, 50V 2.2F Ceramic Capacitor, X5R, Size 0805, 10V 220F Aluminum Capacitor, SMD, 35V 10F Ceramic Capacitor, X5R, Size 1210, 16V 1F Ceramic Capacitor, X5R, Size 0805, 10V 2.2nF Ceramic Capacitor, X7R, Size 0603, 50V 560F OSCON Capacitor, 6.3V 100F Ceramic Capacitor, X5R, Size 1210, 6.3V 1nF Ceramic Capacitor, X7R, Size 0603, 50V Small Signal Schottky Diode 1.0H Inductor, 20A Saturation Current 30V N-Channel MOSFET 7.0m Rds(on) @ 4.5V 2.21 Resistor, Size 0603, 1%
Qty 4 1 1 1 1 1 1 1 1 1 1 2 1
AVX Vishay AVX AVX AVX Sanyo
(3) (2)
AVX AVX Vishay Cooper (4) Bussmann Fairchild
(5)
Vishay Dale
September 2009
34
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164-3 12V to 1.8V @ 10A)
Item R5 R6 R9 R15 U1 U2
Notes: 1. 2. 3. 4. AVX: www.avx.com Vishay: www.vishay.com Sanyo: www.sanyo.com Cooper: www.cooperbussmann.com Fairchild: www.fairchildsemi.com Micrel, Inc: www.micrel.com.
Part Number CRCW06031R21FKEY3 CRCW06031002FKEY3 CRCW06032001FKEY3 CRCW06038061FKEY3 MIC2164-3YMM MIC5233-5.0YM5
Manufacturer Vishay Dale Vishay Dale Vishay Dale Vishay Dale Micrel. Inc.
(6)
Description 1.21 Resistor, Size 0603, 1% 10k Resistor, Size 0603, 1% 2k Resistor, Size 0603, 1% 8.06k Resistor, Size 0603, 1% 1MHz Buck Controller LDO
Qty 1 1 1 1 1 1
Micrel. Inc.
5. 6.
September 2009
35
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Package Information
10-Pin MSOP (MM)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2009 Micrel, Incorporated.
September 2009
36
M9999-090409-B

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