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200 khz, 1 a high voltage step - down switching regulator data sheet adp3050 rev. c information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by ana log devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. specifications subject to change without notice. no license is granted by implication or otherwise under any patent or patent right s of analog devices. trademarks and registered trademarks are the property of their respective owners. one technology way, p.o. box 9106, norwood, ma 02062 - 9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.461 .3113 ? 2008 C 2012 analog devices, inc. all rights reserved. features wide i nput v oltage r ange: 3.6 v to 30 v adjustable and f ixed (3.3 v, 5 v) o utput o ptions integrated 1 a p ower s witch uses s mall s urface - m ount c omponents cycle - b y - c ycle c urrent l imiting peak i nput v oltage (100 ms): 60 v configurable as a b uck, b uck - b oost , and sepic r egulator available in 8 - lead soic package support ed by adisimpower ? design tool applications industrial p ower s ystems pc p eripheral p ower s ystems preregulator for l inear r egulators dist ributed p ower s ystems automotive s ystems battery c hargers functional block dia gram 200khz oscillator frequency and current limit foldback 2.50v regulator current sense amplifier boost fb sd gnd bias current limit g m 1.2v cmp r s q driver switch in comp + adp3050 3 6 7 4 5 1 2 8 00125-001 figure 1 . general description the adp3050 is a current mode monolithic buck (step down) pwm switching regulator that contains a high current 1 a power switch and all control, logic, and protection functions. it uses a unique compensation scheme allowing the use of any type of output capacitor (tantalum, ceramic, electrolytic, os - con). unlike some buck regulators, the design is not restricted to using a specific type of output capacitor or esr value. a special boosted drive stage is used to saturate th e npn power switch, providing a system efficiency higher than conventional bipolar buck switchers. further efficiency improvements are obtained by u sing the low voltage regulated output to provide the internal operating current of the device . a high switching frequency allows the use of small external surface - mount compo - nents. a wide variety of standard off - the - shelf devices can be used, providing a great deal of design flexibility. a complete regulator design requires only a few external components. the adp3050 includes a shutdown input that places the device in a low power mode, reducing the total supply current to under 20 a. internal protection f eatures include thermal shutdown circuit ry and a cycle - by - cycle current limit for the power switch to provide complete device protection under fault conditions. the adp3050 provides excellent line and load regulation, maintaining typically less than 3% ou tput voltage accuracy over temperature and under all input voltage and output current conditions. the adp3050 is specified over the industrial temperature range of ? 40 c to +85 c and is available in a thermally enhanced 8 - lead ( not pb - free only) soic packa ge and a standard 8 - lead ( pb - free only) rohs - compliant soic package .
adp3050 data sheet rev. c | page 2 of 20 table of contents features .............................................................................................. 1 applications ....................................................................................... 1 functional block diagram .............................................................. 1 general description ......................................................................... 1 revision history ............................................................................... 2 specifications ..................................................................................... 3 absolute maximum ratings ............................................................ 4 esd caution .................................................................................. 4 pin configuration and function descriptions ............................. 5 typical performance characteristics ............................................. 6 theory of operation ...................................................................... 10 setting the output voltage ........................................................ 10 applications information .............................................................. 11 adisimpower design tool ....................................................... 11 inductor selection ...................................................................... 11 output capacitor selection ....................................................... 12 catch diode selection ............................................................... 13 input capacitor selection .......................................................... 14 discontinous mode ringing ..................................................... 14 setting the output voltage ........................................................ 14 frequency compensation ......................................................... 14 current limit/freque ncy foldback ......................................... 15 bias pin connection .................................................................. 15 boosted drive stage ................................................................... 15 st art - up/minimum input voltage ........................................... 15 thermal considerations ............................................................ 16 board layout guidelines ............................................................... 17 typical applications ................................................................... 17 inverting (buck boost) regulator ............................................ 18 outline dimensions ....................................................................... 20 ordering guide .......................................................................... 20 revision history 6/12 rev. b to rev. c change to features section ............................................................. 1 added adisimpower design tool section ................................. 11 changes to ordering guide .......................................................... 20 3/08 rev. a to rev. b upda ted format .................................................................. universal changes to general description s ection ...................................... 1 changes to figure 3 and figure 5 ................................................... 6 changes to table 2 ............................................................................ 4 deleted table 4 ................................................................................ 14 changes to tabl e 5 .......................................................................... 15 deleted table 8 ................................................................................ 15 deleted table 7 ................................................................................ 16 deleted table 9 ................................................................................ 16 changes to boosted drive stage s ection a nd thermal considerations s ection s ................................................................. 19 changes to figure 27 ...................................................................... 2 0 changes to ordering guide .......................................................... 23
data sheet adp3050 rev. c | page 3 of 20 specifications v in = 10 v, t a = ? 40 c to +85 c, unless otherwise noted . table 1. parameter 1 symbol conditions min typ max unit feedback feedback voltage v fb over l ine and t emperature adp3050 1.16 1.20 1.24 v adp3050 -3.3 3.20 3.30 3.40 v adp3050 -5 4.85 5.00 5.15 v line regulation v in = 10 v to 30 v, no load 0.005 %/v load regulation i load = 100 ma to 1 a, adp3050ar o nly ? 1.0 +0.1 +1.0 %/a adp3050ar - 3.3, adp3050ar -5 ? 0.5 +0.1 +0.5 %/a input bias current i fb adp305 0ar o nly 0.65 2 a error amplifier transconductance 2 g m 1250 mho voltage gain 2 a vol 300 v/v output current adp3050 comp = 1.0 v, fb = 1.1 v to 1.3 v 115 a adp3050 -3.3 comp = 1.0 v, fb = 3.0 v to 3.6 v 120 a adp3050 - 5 comp = 1.0 v, fb = 4.5 v to 5.5 v 135 a oscillator oscillator frequency 3 f osc 170 200 240 khz minimum duty cycle d min 10 % maximum duty cycle d max 90 % switch average o utput current limit 4 i cl(avg) adp3050 boost = 15 v, fb = 1.1 v 1.0 1.25 1.5 a adp3050 -3.3 boost = 15 v, fb = 3.0 v 1.0 1.25 1.5 a adp3050 -5 boost = 15 v, fb = 4.5 v 1.0 1.25 1.5 a peak switch current limit 5 i cl(peak) 1.5 1.7 2.1 a saturation voltage boost = 15 v, i load = 1 a 0.65 0.95 v leakage current 50 na shutdown input voltage low 0.4 v input voltage high 2.0 v supply input voltage range 6 v in 3.6 30 v minimum bias voltage v bias 3.0 v minimum boost voltage v boost 3.0 v in supply current i q normal mode bias = 5.0 v 0.7 1.5 ma shutdown mode sd = 0 v, v in 30 v 15 40 a bias supply current i bias bias = 5.0 v 4.0 6.0 ma boo st supply current i boost boost = 15 v, i sw = 0.5 a 18 ma boost = 15 v, i sw = 1.0 a 20 40 ma 1 all limits at temperature extremes are guaranteed via correlation using standard s tatistical q uality c ontrol (sqc). 2 transconductance and voltage gain measurements refer to the internal amplifier without the voltage divider. to calculate the transcondu ctance and gain of the fixed voltage parts, divide the values shown by fb/1.20. 3 the switching frequency is reduced when the feedback pin is lower than 0.8 fb. 4 see figure 24 for t ypical application circuit. 5 switch current l imit is measured with no diode, no inductor, and no output capacitor. 6 minimum input voltage is not measured directly, but is guaranteed by other tests. the actual minimum input voltage needed t o keep the output in regulation depend s on output voltage and load current.
adp3050 data sheet rev. c | page 4 of 20 absolute maximum rat ings table 2. parameter rating in voltage continuous ? 0.3 v to +40 v peak (<100 ms) ? 0.3 v to +60 v boost voltage continuous ? 0.3 v to +45 v peak (<100 ms) ? 0.3 v to +65 v sd , bias voltage ? 0.3 v to in + 0.3 v fb voltage ? 0.3 v to +8 v comp voltage ? 0.3 v to in + 0.3 v switch voltage ? 0.3 v to in + 0.3 v operating ambient te mperature range ? 40c to +85c operating junction temperature range ? 40c to +125c storage temperature range ? 65c to +150c ja (4 - layer pcb) 1 60.6 c/w ja (4 - layer pcb) 2 87.5c/w lead temperature ( s oldering, 60 sec) 300c 1 applied to all models that are not pb - free . 2 applied to all pb - free models. stresses above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions a bove those indicated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. esd caution
data sheet adp3050 rev. c | page 5 of 20 pin configuration an d function descripti ons switch 1 boost 2 bias 3 fb 4 in 8 gnd 7 sd 6 com p 5 adp3050 t op view (not to scale) 00125-002 figur e 2 . pin configuration table 3 . pin function descriptions pin no. mnemonic description 1 switch switch node. this pin is the emitter of the internal npn power switch. the voltage at this pin switches betw een v in and approximately ? 0.5 v. 2 boost boost pin. this pin is used to provide a boosted voltage (higher than v in ) for the drive stage of the npn power switch. with the higher drive voltage, the power switch can be saturated, greatly reducing the switch power losses. 3 bia s bias input pin. connect this pin to the regulated output voltage to maximize system efficiency. when this pin is above 2.7 v, most of the adp3050 operating current is taken from the output instead of the input supply. leave unconnected if not used. 4 fb feedback pin. this feedback pin senses the regulated output voltage. conne ct this pin directly to the out put (fixed output versions). 5 comp compensation node. this pin is used to compensate the regulator with an external resistor and capacitor. t his pin is also us ed to override the control loop. howe ver, the voltage on this pin should not exceed 2 v, because the pin is internally clamped to ensure a fast transient response. use a pull - up resistor if this pin is to be pulled higher than 2 v. 6 sd shutdown pin. use this pin to turn the device on and off. if this feature is not needed, tie this pin directly to in . 7 gnd ground pin. connect this pin to local ground plane. 8 in power input. connect this pin to the input supp ly voltage. an input bypass capacitor must be placed close to this pin to ensure proper regulator operation.
adp3050 data sheet rev. c | page 6 of 20 typical performance characteristics temperature (c) v in = 10v, no load 1.5 0 ?45 ?35 quiescent operating current (ma) ?25 ?15 ?5 5 15 25 35 45 55 65 75 85 4.5 2.0 1.0 0.5 3.5 2.5 4.0 3.0 into bias pin into in pin 5.0 00125-003 figure 3 . quiescent operating current vs. temperature 00125-004 supply voltage (v) 25 20 0 30 0 shutdown quiescent current (a) 15 10 5 5 10 15 20 25 figure 4 . shutdown quiescent current vs. supply voltage 00125-005 supply voltage (v) 10 8 0 quiescent operating current (ma) 6 4 2 bias tied to v out v out = 3.3v v out = 5v 30 0 5 10 15 20 25 figure 5 . quiescent operating current vs. supply voltage temperature (c) v in = 10v 0.6 0 average output current (a) 1.8 0.8 0.4 0.2 1.4 1.0 1.6 1.2 2.0 00125-006 ?45 ?35 ?25 ?15 ?5 5 15 25 35 45 55 65 75 85 figure 6 . average output current limit vs. temperature load current (a) 25 20 0 0 0.1 0.2 boost current (ma) 15 10 5 v in = 10v 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 00125-007 figure 7 . boost current vs. load current output current (ma) 100 90 0 0 200 400 600 800 k 1 60 30 20 10 80 70 40 50 efficiency (%) v in = 6v v in = 24v l = 33h c in = 22f c out = 100f v in = 12v 00125-008 v in = 30v v in = 18v figure 8 . 5 v output efficiency
data sheet adp3050 rev. c | page 7 of 20 v in = 12v output current (ma) 100 90 0 1k 0 200 400 600 800 60 30 20 10 80 70 40 50 efficiency (%) l = 33h c in = 22f c out = 100f v in = 5v v in = 18v v in = 24v v n = 30v 00125-009 figure 9 . 3.3 v output efficiency temperature (c) ?0.5 output voltage change (%) ?45 ?35 ?25 ?15 ?5 5 15 25 35 45 55 65 75 85 00125-010 0.4 0.5 ?0.4 ?0.2 ?0.3 ?0.1 0.1 0 0.2 0.3 v in = 10v i load = 1a figure 10 . output voltage change vs. temperature input voltage (v) 0 output voltage change (%) 10 20 30 0.6 0.2 0.4 0 0.6 0.4 0.2 v out = 5v i load = 1a i load = 100ma 00125-0 1 1 fig ure 11 . 5 v output voltage change vs. input voltage input voltage (v) 0 output voltage change (%) 10 20 30 0.6 0.2 0.4 0 0.6 0.4 0.2 v out = 3.3v i load = 1a i load = 100ma 00125-012 figure 12 . 3.3 v output voltage change vs. input voltage load current (a) 8 5 2 minimum input voltage (v) 7 6 4 3 v out = 3.3v v out = 5v 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 00125-013 figure 13 . minimum input voltage vs. load current load current(a) ?0.18 output voltage change (%) ?0.16 ?0.14 ?0.12 ?0.10 ?0.08 ?0.06 ?0.04 ?0.02 0 v in = 10v 00125-014 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 figure 14 . load regulation
adp3050 data sheet rev. c | page 8 of 20 load current (a) 0.8 0 switch saturation voltage (v) 0.1 0.2 0.3 0.4 0.5 0.6 0.7 v in = 10v 00125-015 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 figure 15 . switch saturation voltage vs. load current ambient temperature (c) 196 190 switching frequency (khz) 198 194 192 204 200 208 202 v in = 10v i load = 250a 206 210 ?45 ?35 ?25 ?15 ?5 5 15 25 35 45 55 65 75 85 00125-016 figure 16 . switching frequency vs. temperature normalized feedback voltage (v) 250 200 0 0 1.0 0.2 0.4 0.6 0.8 switching frequency (khz) 150 100 50 v in = 10v comp = 0.4v 00125-017 figure 17 . frequency foldb ack time (1s/div) v in = 10v v out = 5v i load = 800ma l = 33h c in = 22f c out = 100f v sw = 5v/div i l = 500ma/div 0v 0a 00125-018 figure 18 . continuous conduction mode waveforms time (1s/div) v in = 10v v out = 5v i load = 100ma l = 33h c in = 22f c out = 100f v sw = 5v/div i l = 500ma/div 0v 0a 00125-019 figure 19 . discontinuous conduction mode waveforms time (400s/div) v out = 200mv/div i load 5v 1a 0a 00125-020 v in = 10v v out = 5v i load = 100ma to 1a switched l = 33h c in = 22f c out = 100f figure 20 . transient response
data sheet adp3050 rev. c | page 9 of 20 time (100s/div) v out = 1v/div i l = 500ma/div 0v 0a 00125-021 v in = 10v v out = 5v r load = 19? l = 33h coiltronics up2b-330 c in = 22f c out = 100f figure 21 . start - up from shutdown temperature (c) 1150 1000 ?45 ?35 transconductance (mho) ?25 ?15 ?5 5 15 25 35 45 55 65 75 85 1450 1200 1100 1050 1350 1250 1400 1300 1500 00125-022 v in = 10v, no load figure 22 . error amplifier transconductance vs. temperature frequency (hz) 1 1m 100 magnitude (db) 1k 10 k 100k 57.6 48.0 38.4 28.8 19.2 9.6 0 ?28.8 ?19.2 ?9.6 ?38.4 220 200 180 160 140 120 100 40 60 80 20 phase (degrees) no load 00125-023 figure 23 . error amplifier gain
adp3050 data sheet rev. c | page 10 of 20 theory of operation the adp3050 is a fixed frequency, current mode buck re gulator . current mode systems provide excellent transient response, and are much easier to compen sate than voltage mode systems ( r efer to figure 1 ) . at the beginning of each clock cycle, the oscillator sets the latch, turning on the power switch. the signa l at the noninverting input of the comparator is a replica of the switch current (summed with the oscillator ramp). when this signal reaches the appropriate level set by the output of the error amplifier, the comparator resets the latch and turns off the p ower switch. in this manner, the error amplifier sets the correct current trip level to keep the output in regulation. if the error amplifier output increases, more current is delivered to the output; if it decreases, less current is delivered to the outpu t. the current sense amplifier provides a signal proportional to switch current to both the comparator and to a cycle - by - cycle current limit. if the current limit is exceeded, the latch is reset, turning the switch off until the beginning of the next clock cycle. the adp3050 has a foldback current limit that reduces the switching frequency under fault conditions to reduce stress to the ic and to the external components. most of the control circuitry is biased from the 2.5 v internal regulator. when the bias pin is left open, or when the voltage at this pin is less than 2.7 v, all of the operating current for the adp3050 is drawn from the input supply. when the bias pin is above 2.7 v, the majority of the operating current is drawn from this pin (usually tied to the low voltage output of the regulator ) instead of from the higher voltage input supply. this can provide substantial e fficiency improvements at light load conditions, especially for systems where the input voltage is much higher than the output volta ge. the adp3050 uses a special drive stage allowing the power switch to saturate. an external diode and capacitor provide a boosted voltage to the drive stage that is higher than the input supply voltage. overall efficiency is dramatically improved by usin g this type of saturating drive stage. pulling the sd pin below 0.4 v puts the device in a low power mode, shutting off all internal circuitry and reducing the supply current to under 2 0 a. u1 adp3050-3.3 v in c3 220nf d1 1n5818 12v c1 22f + l1 33h v out 3.3v r1 4k? c2 1nf + c4 100f d2 1n4148 1 2 3 4 switch boost bias fb in gnd sd comp 8 7 6 5 00125-024 figure 24 . typical application circuit setting the output v oltage the output of the adjustable version (adp3050ar and ADP3050ARZ ) can be set to any voltage between 1.25 v and 12 v by connecting a resistor divider to the fb pin as shown in figure 25 . ? ? ? ? ? ? ? ? ? = 1 2 . 1 out v r1 r2 (1) u1 adp3050 v in r1 20k? r2 21.5k? c f c3 0.22f d1 1n5817 gnd 5v c1 210f ceramic + c2 0.01f l1 22h v out 2.5v r c 7.5k? c c 4.7nf d2 1n4148 + c4 222f ceramic 1 2 3 4 8 7 6 5 00125-025 switch boost bias fb in gnd sd comp figure 25 . adjustable output application circuit
data sheet adp3050 rev. c | page 11 of 20 a pplication s i nformation adi sim p ower design tool the adp 3050 is supported by th e adisimpower design tool set. adisimpower is a collection of tools that produce complete power designs optimized for a specific design goal. the tools en able the user to generate a full schematic, bill of materials, and calculate performance in minutes. adisimpower can optimize designs for cost, area, efficiency, and parts count while taking into considera - tion the operating conditions and limitations of the ic and all real external components. for more information about adisimpower design tools, refer to www.analo g.com/adisimpower . the tool set is available from this website, and users can request an unpopulated board through the tool. the complete process for designing a step - down switching regulator using the adp3050 is provided in the following sections. each s ection includes a list of recommended devices. these lists do not include every available device or manufacturer. they con tain only surface - mount devices. e quivalent through - hole devices can be substituted if needed. in choosing components , keep in mind wh at i s most important to the design, for example, efficiency, cost, and size. t hese ultimately determine which compo - nents are used . it is also important to ensure that the design specifications are clearly defined and reflect the worst - case conditions. key specifications include the minimum and maximum input voltage, the output voltage and ripple, and the minimum and maximum load current. i nductor s election the inductor value determine s the mode of operation for the regulator: continuous mode, where the ind uctor current flows continuously; or discontinuous mode, where the inductor current reduces to zero during every switch cycle. continuous mode is the best choice for many applications. it provides higher output power, lower peak currents in the switch, ind uctor , and diode, and a lower inductor ripple current, which me ans lower output ripple voltage . discontinuous mode allow s the use of smaller magnetics, but at a pric e: lower available load current and higher peak and ripple currents. designs with a high in put voltage or a low load current often operate in discontinuous mode to minimize inductor value and size. the adp3050 is designed to work well in both modes of operation. continuous mode the inductor current in a continuous mode system is a triangular wav eform (equal to the ripple current) centered around a dc value (equal to the load current). the amount of ripple current is determined by the inductor value, and is usually between 20% and 40% of the maximum load current. to reduce the inductor size, rippl e currents between 40% and 80% are often used in continuous mode designs with a high input voltage or a low output current. the inductor value is calculated using the following equation: ) ( ) ( 1 max in out sw ripple out max in v v f i v v l ? = (2) where v in(max) is the maximum input v oltage, v out is the regulated output voltage, and f sw is the switching frequency (200 khz). the initial choice for the amount of ripple current may seem arbitrary, but it serve s as a good starting point for finding a standa rd off - the - shelf inductor value, such as 10 h, 15 h, 22 h, 33 h, and 47 h . if a specific inductance value is to be used, simply rearrange e quation 2 to find the ripple current. for an 800 ma, 12 v to 5 v system, and a ripple current of 320 ma (40% of 800 ma) is chosen, the inductanc e is h 45.5 12 5 10 200 1 0.32 5 12 3 = ? = l a 47 h inducto r is the closest standard value that gives a ripple current of about 310 ma. the peak switch current is equal to the load current plus one - half the ripple current (this is also the peak current for th e inductor a nd the catch diode) . a 95 . 0 155 . 0 8 . 0 2 1 ) ( ) ( = + = + = ripple max out pk sw i i i (3) pick an inductor with a dc (or saturation) current rating about 20% larger than i sw(pk) to ensure that the inductor is not running near the edge of saturation. for this example, 1.20 0.95 a = 1.14 a , use a n inductor with a dc current rating of at least 1.2 a. the maxi - mum switch current is internally limited to 1.5 a, and this limit, along with the ripple current, determine s the maximum load current the system can provide. if the load current decreases to b elow one - half the ripple current, the regulator operate s in discontinuous mode. discontinuous mode for load currents less than a pproximately 0.5 a, discontinuous mode operation can be used. this allow s the use of a smaller inductor, but the ripple current is much higher (which means a higher output ripple voltage). if a larger output capacitor must be used to reduce the output ripple voltage, the overall system may take up more board area than if a larger inductor i s used. the operation and equations for th e two modes are quite different, but the boundary between these two modes occurs when the ripple current is equal to twice the load current (when i ripple = 2 i out ). from this , e quation 2 is used to find the minimum inductor value needed to keep the syste m in continuous mode operation ( solve for the inductor value with i ripple = 2 i out ). ) ( ) ( 1 2 max in out sw out out max in dis v v f i v v l ? = (4) using an inductor below this value cause s the system to operate in discontinuous mode.
adp3050 data sheet rev. c | page 12 of 20 f or a 400 ma, 24 v to 5 v system h 7 . 24 24 5 10 200 1 4 . 0 2 5 24 3 ? dis l if the chosen inductor value is too small, the internal current limit trip s each cycle and the regulator has trouble providing the necessary load current. inductor core types and materials many types of inductors are currently available. numerous core styles along with numerous core materials often make the selection process seem even more confusing. a quick overview of the types of inductors available make s the selection process a little easier to understand. open core geometries (bobbin core) are usu ally less expensive than closed core geometries (toroidal core) and are a good choice for some applications, but care must be taken when they are used. in open core inductors, the magnetic flux is not completely contained inside the core. the radiating mag netic field generate s e lectro m agnetic i nterference (emi), often inducing voltages onto nearby circuit board traces. these inductors may not be suitable for systems that contain very high accuracy circuits or sensitive magne tics. a few manufacturers have se miclosed and shielded cores, where an outer magnetic shield surrounds a bobbin core. these devices have less emi than the standard open core and are usually smaller than a closed core. most core materials used in surface - mount inductors are either powdered iron or ferrite. for many designs, material choice is arbitrary, but the properties of each material should be recognized . ferrites have lower core losses than powdered iron, but the lower loss means a higher price. powdered iron cores saturate softly (th e inductance gradually reduces as current rating is exceeded), wh ereas ferrite cores saturate much more abruptly (the inductance rapidly reduces). kool m ? is one type of ferrite that is specially designed to minimize core losses and heat generation (espec ially at switching frequencies above 100 khz), but again, these devices are more expensive. t he winding dc resistance (dcr) of the inductor must not be overlook ed . a high dcr can decrease system efficiency by 2% to 5% for lower output voltages at heavy loa ds. to obtain a lower dcr means using a physically larger inductor, so a trade - off in size and efficiency must be made. the power loss due to this resistance is i out 2 dcr. for an 800 ma, 5 v to 3.3 v system with an inductor dcr of 100 m, the winding resistance dissipate s (0.82 a) 2 0.1 = 64 mw. this represents a power loss to the system of 64 mw/(3.3 v 800 ma) = 2.4%. typical dcr values are between 10 m and 200 m. choosing an inductor several considerations must be made when choosing an inductor : cost, size, emi, core and copper losses, and maximum current rating. use the following steps to choose an inductor that is right for the system (refer to the calculations and descriptions in the i nductor s election section ). contact the manufacturers for their full product offering, availability , and pricing. the manufacturers offer many more values and package sizes to suit numerous applications. 1. choose a mode of operation, the n calculate the inductor value using the appropriate equation. for continuous mode systems, a ripple current of 40% of the maximum load current is a good starting point. the inductor value can then be increased or decreased , if desired. 2. calculate the peak switch current (this is the maximum current seen by the inductor). make sure that the dc (or saturation) current rating of the indu ctor is high enough (around 1.2 the peak switch current). inductors with dc current ratings of at least 1 a should be used f or all designs. this provide s a safety margin for start - up and fault conditions where the inductor current is higher than normal. if the current rating of an inductor is exceeded, the core saturate s , causing the inductance value to decrease and the tempera ture of the inductor to increase. 3. estimate the dc winding resistance based on the inductance value. a general rule is to allow approximately 5 m of resistance per h of inductance. 4. pick the core material and type. first , decide if an open - core inductor can be used with the design. if this cannot be determined , try a few samples of each type (open core, semi closed core, shielded core, and clo sed core). do not be discouraged from using open core inductors because th ey require extra care; just be aware of what to look for if used . they are quite small and inexpensive, and are used successfully in many different applications. o utput c apacitor s el ection the adp3050 can be used with any type of output capacitor. the trade - offs between price, component size, and regulator performance can be evaluated to determine the best choice for each application. the e ffective s eries r esistance (esr) of t he capac itor plays an important role in both the loop compensation and the system performance. the esr provides a 0 in the feedback loop ; therefore , the esr value must be known so the loop can be compensated correctly (most manufacturers specify maximum esr in the ir data sheets). the capacitor esr also contributes to the output ripple voltage (v ripple = esr i ripple ). solid tantalum or multilayer ceramic capacitors are recommended , providing good performance with a small size and reasonable cost . solid tantalum ca pacitors have a good combination of low esr and high capacitance, and are available from several different manufacturers. capacitance values from 22 f to more than 500 f can be used, but values of 47 f to 220 f are sufficient for most designs. a smalle r value can be used, but esr is size - dependent, so a smaller device has a higher esr. ensure that the ripple current of the capacitor rating is larger than the inductor ripple current (the ripple current flow s into the output capacitor). multilayer ceramic capacitors can be used in applications where minimum output voltage ripple is a priority. they have a very
data sheet adp3050 rev. c | page 13 of 20 low esr (a 22 f ceramic can have an esr one - fifth that of a 22 f solid tantalum), but may require more board area for the same value of output ca pacitance. a few manufacturers have recently improved upon their low voltage ceramic capacitors, providing a smaller package with a lower esr ( nec tokin, murata, taiyo yuden , and avx). several ceramics can be used in parallel to give an extremely low esr a nd a good value of capacitan ce. if the design is cost sensitive and not severely space limited , several aluminum electrolytic capacitors can be used in parallel (their size and esr are larger than ceramic and solid tantalum). os - con capacitors can also be used, but they are typically larger and more expensive than ceramic or solid tantalum capacitors. choosing an output capacitor use the following steps to choose an appropriate capacitor . 1. decide the maximum output ripple voltage for the design, and this det ermine s your maximum esr (remember that v ripple esr i ripple ). typical output ripple voltages range between 0.5% and 2% of the output voltage. to lower the output voltage ripple, there are only two choices: either increase the inductor value, or use an output capacitor with a lower esr. 2. decide what type of capacitor to use (tantalum, ceramic, or others ). many more values, sizes, and voltage ratings are available, so contact each manufacturer for a complete product list. if a certain type of capacitor must be used and space permits, use several device s in parallel to reduce the total esr. 3. check the capacitor voltage rating and ripple current rating to ensure it work s for the application in question. t hese ratings are derated for higher temperatures, so always check the manufacturers data sheet . 4. make s ure the final choice for the output capacitor has been optimized for cost, size, availability, and performance yet still meet s the required capacitance. the recom mended capacitance is in the 47 f to 220 f range. c atch d iode s election the recommended cat ch dio de is a type 1n5818 schottky or equivalent. the low forward voltage drop (450 mv typical at 1 a) and fast switching speed of a schottky rectifier provide the best performance and efficiency. the 1n5818 is rated at 30 v reverse voltage and 1 a averag e forward current. for lower input voltages, use a lower voltage schottky to reduce the diode forward voltage drop and incr ease overall system efficiency; for example , a 12 v to 5 v system does not need a 30 v diode . for automotive applications, a 60 v sch ottky may be necessary. the average forward current for the ca tch diode is calculated by in out in out avg diode v v v i i ? = ) ( (5) for the earlier continuous mode example (12 v to 5 v at 800 ma), the average diode current is a 47 . 0 12 5 12 8 . 0 ) ( = ? = avg diode i (6) for this system, a 1n5817 is a good choice (rated at 20 v and 1 a). do not use catch diodes rated less than 1 a. even though the average current can be less than 1 a under normal operating conditions, as the diode current is much higher under fault conditions. the worst - c ase fault condition for the diode occurs when the regulator becomes slightly overloaded (sometimes called a soft short). this is usually only a problem when the input voltage to output voltage ratio is greater than 2.5. under this condition, the load curre nt needed is slightly more than the regulator can provide. the output voltage droops slightly, and the switch stays on every cycle until the internal current limit is reached. under this condition, the load current can reach around 1.2 a. for examp le, when using a system with an input voltage of 24 v and an output voltage of 5 v , if a gradual overload causes the output voltage to droop to 4 v, the average diode current is a 0 . 1 24 4 24 2 . 1 ) ( = ? = avg diode i (7) if the system must survive such gradual overloads for a pr olonged period of time, ensure the diode chosen can survive these conditions. a larger 2 a or 3 a diode can be used if necessary. table 4. manufacturers inductor manufacturers capacitor manufacturers schottky diode manufacturers s umida avx motorola coilcraft kemet diodes, inc. cooper bussmann coiltronics murata international rectifier nec tokin nemco nihon inter electronic s w rth elektronik vishay sprague toko nec tokin taiyo yuden
adp3050 data sheet rev. c | page 14 of 20 choosing a catch di ode use the following steps to pick an appropriate catch diode. table 5 shows several schottky rectifiers with different reverse voltage and forward current ratings. the average diode current rating must be sufficient to provide the required load current (see the calculations in the previous section ). diodes rated below 1 a should not be used, even if the average diode current is much lower. the reverse voltage rating of the catch diode should be at least the maximum input voltag e. often a higher rating is chosen (1.2 the maximum input voltage) to provide a safety margin . table 5. schottky diode selection guide v r 1 a 2 a 3 a 15 v 10bq15 30bq15 20 v 1n5817 b220 sk32 30 v v1n5818 b230 sk33 40 v 1n5819 b240 sk34 i nput c apacitor s election the input bypass capacitor plays an important role in proper regulator operation, minimizing voltage transients at the input and providing a short local loop for the switching current. place this capacit o r close to the adp3050 between the in and gnd pins using short, wide traces. this input capacitor should have an rms ripple current rating of at least 2 ) ( ? ? ? ? ? ? ? ? ? (8 ) this rating is crucial because the input capacitor must be able to withstand the large current pulses present at the input of a step - down regulator. values of 20 f to 50 f are typical, but the main criteria for capacitor selection is the ripple current and voltage ratings. ceramics are an excellent choice for input bypassing , du e to their low esr and high ripple current rating. ceramics are especially suited for high input voltages and are available from many different manufacturers. tantalums are often used for input bypassing , but precautions must be taken because they occasion ally fail when subjected to large inrush currents during power - up. these surges are common when the regulator input is connected to a battery or high capacitance supply. several manufacturers now offer surface - mount solid tantalum capacitors that are surge tested, but even these devices can fail if the current surge occurs when the capacitor voltage is near its maximum rating. for this reason, a 2:1 derating is suggested for tantalum capacitors used in applications where la r ge inrush currents are present. f or example , a 20 v tantalum should be used only for an input voltage up to 10 v . aluminum electrolytics are the cheapest choice, but it takes several in parallel to get a good rms current rating. os - con capacitors have a good esr and ripple current rating, but they are typically larger and more costly. refer to table 4 for a list of capacitor manufacturers . d iscontinous m ode r inging when operating in discontinuous mode, high frequency ringing appear s at the switch n ode when the inductor current has decreased t o zero. this ringing is normal and is not a result of loop instability. it is caused by the switch and diode capacitance reacting with the inductor to form a damped sinusoidal ringing. this ringing is usually in the range of several megahertz, and is not harmful to normal circuit operation. s etting the o utput v oltage the fixed voltage versions of the adp3050 (3.3 v and 5 v) have the feedback resistor divider included on - chip. for the adjustable version, the outpu t voltage is set using two external resistors. referring to figure 25 , pick a value for r1 between 10 k and 20 k, then calculate the appropriate value for r2 using the following equation : ? ? ? ? ? ? ? = 1 20 . 1 out v r1 r2 ( 9 ) it is important to note that the accuracy of these resistors directly affects the accuracy of the output voltage. th e fb pin threshold variation is 3%, and the tolerances of r1 and r2 add to this to determine the total output variation. use 1% resistors placed close to the fb pin to prevent noise pickup. f requency c ompensation the adp3050 uses a unique compensation scheme that allows the use of any type of output capacitor. the designer is not limited t o a specific type of capacitor or a specific esr range. external compensation allows the designer to optimize the loop for transient response and system performance. t he values for r c and c c set the pole and zero locations for the error amplifier to compensate the regulator loop. for tantalum output capacitors, the typical system compensation values are r c = 4 k and c c = 1 nf; for ceramics, the typical values are r c = 4 k and c c = 4.7 nf. these values may not be optimized for all designs, but they provide a good starting point for selecting the final compensation values. other types of output capacitors require different values of c c between 0.5 nf and 10 nf . typically , the lower the esr of the output capacitor, the larger the value for c c . normal variations in capacitor esr, output capacitance, and inductor value (due to production tolerances, changes in operating point, changes in temperature) affect the loop gain and phase response. always check the final design over its complete operating range to ensure proper regulator operation. adjusting the r c and c c values can optimize compensation. use the typical values above as a starting point, then try increasing and decre asing each independently and observing the transient response. an easy way to check the transient response of the design is to observ e the output while pulsing the load current at a rate of approximately 100 hz to 1 khz. there should be some slight ringing at the output when the load pulses, but this should not be excessive (just a few rings). the frequency of this ringing
data sheet adp3050 rev. c | page 15 of 20 shows the approximate unity - gain frequency of the loop. again, always check the design over its full operating range of input voltage, o utput current, and temperature to ensure that the loop is compensated correctly. in addition to setting the zero location, r c also sets the high frequency gain of the error amplifier. if this gain is too large, output ripple voltage appear s at the comp pin (the output of the error amplifier) with enough amplitude to interfere with norma l regulator operation. if this occurs , subharmonic switching results (the pulse width of the switch waveform change s , even though the output voltage stays regulated). the vol tage ripple at the comp pin should be kept below 100 mv to prevent subharmonic switching from occurring. the amount of ripple can be estimated by the following formula, where g m is the error amplifier transconductance (g m = 1 250 mho): ( ) ( ) out fb ripple c m ripple comp v v esr i r g v = , ( 10) for example, a 12 v to 5 v, 800 ma regulator with an inductor of l = 47 h has i ripple = 310 ma ( see example from the continuous mode section) if a 100 f tantalum output capacitor with a maximum esr of 1 00 m and compensation values of r c = 4 k and c c = 1 nf are used. the ripple voltage at the comp pin is ( ) ( ) mv 2 . 37 0 . 5 20 . 1 1 . 0 310 . 0 10 4 10 1250 3 6 , = = ? ripple comp v (11 ) if this ripple voltage is more than 100 mv, r c need s to be decreased to prevent subharmonic switching. typical values for r c are in the range of 2 k to 10 k. for output voltages greater than 5 v, it may be necessary to add a small capacitor in parallel with r2, as shown in figure 25. this improve s stability and transient response. for tantalum outp ut capacitors, the typical value for c f is 100 pf. for ceramic output capacitors, the typical value for c f is 400 pf. c urrent l imit /f requency f oldback the adp3050 uses a cycle - by - cycle current limit to protect the device under fault and high stress conditi ons. when the current limit is exceeded, the power switch turns off until the beginning of the next oscillator cycle. if the voltage on the feedback pin drops below 80% of its nominal value, the oscillator frequency starts to decrease (see figure 17 in th e typical performanc e characteristics section). the frequency gradually reduces to a minimum value of approximately 80 khz (this minimum occurs when the feedback voltage falls to 30% of its nominal value). this reduces the power di ssipation in the ic, the external diode, and the inductor during short - circuit conditions. this frequency foldback method provides complete device fault protection without interfering with the normal device operation. b ias p in c onnection to help improve e fficiency, most of the internal operating current can be drawn from the lower voltage regulated output voltage instead of the input supply. for example, if the input voltage is 24 v and the output voltage is 5 v, a quiescent current of 4 ma waste s 96 mw if drawn from the input supply, but only 20 mw is drawn from the regulated 5 v output. this power savings is most evident at high input voltages and low load currents. the output voltage must be 3 v or higher to take advantage of this feature. b oosted d rive s tage an external capacitor and diode are used to provide the boosted voltage needed for the special drive stage. if the output voltage is above 4 v, connect the anode of the boost diode to the regulated output; for output voltages less than or equal to vo ltages of 3 v, connect it to the input supply. for some low voltage systems , such as 5 v to 3.3 v converters , the anode of the boost diode can be connected to either the input or output voltage. during switch off time, the boost capacitor is charged up to the volta ge at the anode of the boost diode. when the switch turns on, this voltage is added to the switch voltage (the boost diode is reverse - biased) , providing a voltage higher than the input supply. the peak voltage appearing on the boost pin is the sum of the i nput voltage and the boost voltage (either v in + v out or 2 v in ) . ensure that this peak voltage does not exceed the boost pin maximum rating of 45 v. for most applications, a 1n4148 or 1n914 type diode can be used with a 220 nf capacitor. a 470 nf capacit or may be needed for output voltages between 3 v and 4 v. the boost capacitor should have an esr of less than 2 to ensure that it is adequate ly charged up during switch off time. almost any type of film or ceramic capacitor can be used. s tart -u p /m inimum i nput v oltage for most designs, the regulated output voltage provides the boosted voltage for the drive stage. during startup, the output voltage is 0 , so there is no boosted supply for the drive stage. to deal with this problem, the adp3050 contains a bac kup drive stage to get everything started. as the output voltage increases, so does the boost voltage. when the boost voltage reaches a pprox - imately 2.5 v, the switch drives transition smoothly from the backup driver to the boosted driver. if the boost vol tage d ecrease s below a pproximately 2.5 v , resulting in a short - circuit or overload condition , the backup stage take s over to provide switch drive. the minimum input voltage needed for the adp3050 to function correctly is about 3.6 v (this ensure s proper op eration of the internal circuitry), but a small amount of headroom is needed for all step - down regulators . the following formula gives the approximate minimum input voltage needed for a given system, where v sat is the switch saturation voltage (see figure 15 for the appropriate value of v sat ). figure 13 also shows the typical minimum input voltage needed for 3.3 v and 5 v systems.
adp3050 data sheet rev. c | page 16 of 20 85 . 0 ) ( sat out min in v v v + = ( 12) t hermal c onsiderations several factors contribute to ic power dissipation: ac and dc switch losses, boost current, and quiescent current. the following formulas are used to calculate these losses to determine the power dissipation of the ic. these formulas assume continuous mode operati on, but they provide a reasonable estimate for disconti - nuous mode systems (do not use these formulas to calculate efficiency at light loads). switch loss ( ) sw in out ov in out sat out sw f v i t v v v i p + ? ? ? ? ? ? ? ? = (1 3 ) boost current loss in out sw out boost v v i p 2 = (1 4 ) quiescent current loss ( ) ( ) bias out q in q i v i v p + = (1 5 ) w here : v sat is ~0.6 v at i out = 800 ma (taken from figure 15). f sw is the switch frequency (200 khz). t ov is the switch current /voltage overlap time (~50 ns). sw is the current gain of the npn power switch (~50). i q is the quiescent current drawn from v in (~1 ma). i bias is the quiescent current drawn from v out (~4 ma). for example, a 5 v to 3.3 v system with i out = 800 ma ( ) mw 357 10 200 0 . 5 8 . 0 10 50 0 . 5 3 . 3 6 . 0 8 . 0 3 9 = + ? ? ? ? ? ? = ? sw p mw 35 0 . 5 3 . 3 50 8 . 0 2 = = boost p ( ) ( ) mw 18 10 4 3 . 3 10 5 3 3 = + = ? ? q p for a total ic power dissipation of mw 410 = + + = q boost sw total p p p p (1 6 ) the adp3050 is offered in a thermally enhanced (not pb - free ) 8 - lead soic package with a thermal resistance, ja , of 60.6 c/w , and in a standard pb - free 8 - lead s oic package with ja of 87.5 c / w. the maximum die temperature , t j , is calculated using the thermal resistance and the maximum ambient temperature total ja a j p t t + = (1 7 ) for the previous example (5 v to 3.3 v at 800 ma system, pb - free 8 - lead soic pack age using good layout techniques) with a worst - case ambient temperature of 70 c t j = 70 c + 87.5 c/w 0.41 = 105.9 c the maximum operating junction (die) temperature is 125c, therefore this system operate s within the safe limits of the adp3050. check the die temperature at minimum and maximum supply voltages to ensure proper operati on under all conditions. although t he pcb and its copper traces provide sufficient heat sinking, it is important to follow the layout suggestions in the board layout guidelines section. for any design that combines high output current with high duty cycle and/or high input voltage, the junction temperature must be calculated to ensure normal operation. always use the equations in this section to estimate the power dissipation.
data sheet adp3050 rev. c | page 17 of 20 b oard l ayout g uidelines a good board layout is essential when designing a switching regulator. the high switching currents along with parasitic wiring inductances can generate significant voltage transients and cause havoc in sensitive circuits. for best results, keep the main switching path as tight as possible (keep l1, d1, c in , and c out close together) and minimize the copper area of the switch and boost nodes (without violating current density require - ments) to reduce the amount of noise coupling into other sensitive nodes. adp3050 gnd in switch c in v in gnd d1 c out l1 v out gnd 00125-026 figure 26 . main switching path the external components should be located as close to the adp3050 as possible. for best thermal performance, use wide copper traces for all ic connections, and always connect the gnd pin to a large piece of copper or ground plane. the additional copper i mprove s heat transfer from the ic, greatly reducing the package thermal resistance. further improvements of the thermal performance can be made by using multilayer boards and u sing vias to transfer heat to the other layers. a single layer board layout is shown in figure 27 . the amount of copper used for the input, output, and ground traces can be reduced, but were made large to improve the thermal perfo rmance. for the 5 v and 3.3 v versions, leave out r1 and r2; for the a djustable version, remove the trace that shorts out r2. route all sensitive traces and compo - nents , such as those associated with feedback and compensation , away from the boost and switc h traces. t ypical applications 5 v to 3.3 v buck ( step - d own ) regulator the circuit in figure 28 shows the adp3050 in a buck configuration. it is used to generate 3.3 v regulated output from 5 v input voltage with the following spe cifications: v in = 4.5 v to 5.5 v v out = 3.3 v i out = 0.75 a i ripple = 0.4 a 0.75 a = 0.3 a v out _ ripple = 50 mv output ground input c1 l1 c3 d1 d2 r2 r1 cc rc c2 adp3050 00125-027 figure 27 . recommended board layout u1 adp3050-3.3 v in c3 0.22f d1 1n5817 gnd 5v c1 22f + c2 0.01f l1 22h v out 3.3v r1 7.5k? c4 1nf d2 1n4148 + c5 100f sd 1 2 3 4 8 7 6 5 00125-028 switch boost bias fb in gnd sd comp figure 28 . 5 v to 3.3 v buck regulator
adp3050 data sheet rev. c | page 18 of 20 i nverting ( b uck b oost ) r egulator the circuit in figure 29 shows the adp3050 in a buck - boost configuration that produces a negative output voltage from a positive input voltage. this topology looks quite similar to the buck shown in figure 28 (except the ic and the output filter are now referenced to the negative output instead of ground), but its operation is quite different. for this topology, the feedback pin is grounded and the gnd pin is tied to the negat ive output, allowing the feedback network of the ic to regulate the negative output voltage. 00125-029 u1 adp3050-5 v in d1 1n5818 gnd 12v c1 22f c3 0.22f + c2 0.01f r1 5.1k? c4 3.3nf d2 1n4148 + c5 100f sd ?5v at 0.5a v out l1 47h 1 2 3 4 8 7 6 5 switch boost bias fb in gnd sd comp figure 29 . inverting (buck - boost) regulator the design procedure used for the standard buck converter cannot be used for a buck - boost co nverter due to fundamental differences in how the output voltage is generated. the switch currents in the buck - boost are much higher than the standard buck converter, thus lowering the available load current. to calculate the maximum output current for a g iven maximum switch current , use the following equation: ( ) ? ? ? ? ? ? ? ? + ? + = out in sw out in max sw out in in max out v v l f v v i v v v i 2 ) ( ) ( (1 8 ) where i sw(max) is the switch current limit rating of the adp3050 , and v in is the minimum input voltage. the inductor ripple current is estimated using the following equat ion: out max in out sw max in ripple v v v f l v i + = ) ( ) ( 1 (1 9 ) for the circuit in figure 29 , the maximum ripple current (a t the maximum input voltage) is a 375 . 0 5 12 5 10 200 1 10 47 12 3 6 = ? + ? = ? ripple i high ripple currents are present in both the input and output capacitors, and th eir ripple current ratings must be large enough to sustain the large switching currents present in this topology. the capacitors should have a ripple current rating of at least in out out c c rms v v i i out in ) , ( (20 ) the peak current seen by the diode, switch, and in ductor is found by rearranging the load current equation ? ? ? ? ? ? + ? ? ? ? ? ? ? ? + = ripple out in out in peak i i v v v i 2 1 ( 21) the largest peak currents occur at the lowest input voltage. for this desig n with a load current of 500 ma a 9 . 0 375 . 0 2 1 5 . 0 12 5 12 = ? ? ? ? ? ? + ? ? ? ? ? ? ? ? ? + = peak i (22) the average current diode is equal to the load current. an inductor with a current rating 20% greater than 0.9 a should be used ( a rating of at least 1.2 a). inductors and diodes with ratings greater than 1 a should always be used, even if the calculated peak and average currents are lowe r. this ensure s that start - up and fault conditions do not overstress the components. for the buck - boost topology, the input voltage can be less than the output voltage, such as v in = 4 v or v out = ? 5 v , but the available load current is even lower. the equ ations given in this section are valid for input voltages less than and greater than the output voltage. the voltage seen by the adp3050 is equal to the sum of the input and output voltages (the boost pin sees the sum of v in + 2 |v out |). it is important to en sure that the maximum voltage rating of these pins is not exceeded.
data sheet adp3050 rev. c | page 19 of 20 dual output sepic regulator for many systems, a dual polarity supply is needed. the circuit in figure 30 generates both a positive and a negative 5 v output using a single magnetic component. the two inductors shown are actually two separate windings on a single core contained in a small, surface - mount package. the windings can be connected in parallel or in series to be used as a single inductor for a conven - tional buck regulator, or they can be used as a 1:1 transformer, as in this application. the first winding is used as the standard buck inductor for the +5 v output. the second winding is used to generate the ? 5 v output along with d2, c6, and c7. 00125-030 c1 22f c2 0.01f 1 2 3 4 8 7 6 5 u1 adp3050-5 v in c3 0.22f d1 1n5818 gnd 12v + l1* 25h v out +5v at 0.5a r1 5.1k? c4 1nf d3 1n4148 + c5 100f sd *inductor is a single core with two windings coiltronics ctx25-4 ?5v at 0.25a v out + c6 100f l1* 25h + c7 100f d2 1n5818 switch boost bias fb in gnd sd comp figure 30 . dual output +5 v and ? 5 v regulator these components form a s ingle - e nded p rimary i nductance c onverter (sepic) using the 1:1 coupled inductor to generate the negative supply. when the switch is off, the voltage across the buck winding is equal to v o + v d (v d is the diode drop). this voltage is generated across the second winding, which is connected to produce the ? 5 v supply. the ? 5 v output is generated even without c6 in the circu it, but its inclusion greatly improves the regulation of the negative output and lowers the inductor ripple current. the total output current available for both supplies is limited by the adp3050 (internally limited to around 1.0 a). keeping load currents below 500 ma and 250 ma, for the positive and negative supplies, respectively, ensure s that th e current limit is not reached under normal operation. these limits are not interchangeable; 500 ma cannot be drawn from the ? 5 v supply while drawing only 250 ma from the +5 v supply. the maximum current available from the ? 5 v output is directly related to the +5 v load current , due to the fact that the +5 v output is used to regulate both supplies. typically, the ? 5 v load cu rrent should be around one - half of the +5 v load current to ensure good regulation of both outputs. additionally, the ? 5 v output should have a preload (the minimum current level) of 1% to 2% of the +5 v load current. this help s maintain good regulation of the ? 5 v output at light loads. the ripple voltage of the +5 v output is t hat of a normal buck regulator as described in the applications information section . this ripple voltage is determined by the inductor ripple current and the esr of the output capac itor. for figure 30 , the positive output voltage ripple is a 30 mv peak - to - peak triangular wave. the ripple voltage of the ? 5 v output is a rectangular wave, due to the rectangular shape of the current waveform into the ? 5 v output capacitor . the amplitude of this current waveform is approximately equal to twice the ? 5 v load current. for a load current of 200 ma and an esr of 100 m, the negative output voltage ripple is a pproximately 2 200 ma 100 m, or about 40 mv. the edges of this ripple waveform a re quite fast . a long with the inductance of the output capacitor, it generates narrow spikes on the negative output voltage. these spikes can easily be filtered out using an additional 5 f to 10 f bypass capacitor close to the load (the induct ance of the pcb trace and the additional capacitor create a low - pass filter to remove these high frequency spikes).
adp3050 data sheet rev. c | page 20 of 20 outline dimensions controlling dimensions are in millimeters; inch dimensions (in parentheses) are rounded-off millimeter equivalents for reference only and are not appropriate for use in design. compliant to jedec standards ms-012-aa 012407-a 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) 0.50 (0.0196) 0.25 (0.0099) 45 8 0 1.75 (0.0688) 1.35 (0.0532) seating plane 0.25 (0.0098) 0.10 (0.0040) 4 1 85 5.00 (0.1968) 4.80 (0.1890) 4.00 (0.1574) 3.80 (0.1497) 1.27 (0.0500) bsc 6.20 (0.2441) 5.80 (0.2284) 0.51 (0.0201) 0.31 (0.0122) coplanarity 0.10 figure 31. 8-lead standard small outline package [soic_n] narrow body (r-8) dimensions shown in millimeters and (inches) ordering guide model 1 output voltage temperature range 2 package description package option ordering quantity ADP3050ARZ adj ?40c to +85c 8-lead soic_n r-8 98 ADP3050ARZ-rl adj ?40c to +85c 8-lead soic_n r-8 2,500 ADP3050ARZ-r7 adj ?40c to +85c 8-lead soic_n r-8 1,000 ADP3050ARZ-3.3 3.3 v ?40c to +85c 8-lead soic_n r-8 98 ADP3050ARZ-3.3-rl 3.3 v ?40c to +85c 8-lead soic_n r-8 2,500 ADP3050ARZ-3.3-rl7 3.3 v ?40c to +85c 8-lead soic_n r-8 1,000 ADP3050ARZ-5 5 v ?40c to +85c 8-lead soic_n r-8 98 ADP3050ARZ-5-reel 5 v ?40c to +85c 8-lead soic_n r-8 2,500 ADP3050ARZ-5-reel7 5 v ?40c to +85c 8-lead soic_n r-8 1,000 adp3050-evalz evaluation board 1 z = rohs compliant part. 2 operating junction temperature is ?40 to +125c. ?2008C2012 analog devices, inc. all rights reserved. trademarks and registered trademarks are the property of their respective owners. d00125-0-6/12(c)

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